Battery assisted RFID system command set

ABSTRACT

Specialized battery assisted command set design methods are disclosed that provide for interference rejection using highly sensitive but relatively broadband RFID tags. The command set design also supports RFID system RF power control for further interference control. The command set design also allows for convenient expansion to active transmitters and receivers in tags operating within the same system. Embodiments of the present invention provide RFID systems having battery-assisted, Semi-Passive RFID tags that operate with sensitive transistor based square law tag receivers utilizing a plurality of tag receiver dynamic range states. Embodiments of the present invention are also enhanced with receiver training and synchronizing methods suited to the high tag sensitivity and need for dynamic range state switching. These enhancements may employ pseudo-random sequence based receiver training, activation signaling, and frame synchronizing. Additional enhancement attained via power leveling methods that optimize the amount of transmitted power and interference from a reader in relation to the sensitivity of the RFID tags, their ranges from the reader, and the unique physics of the backscatter RFID radio link.

BACKGROUND

A. Field of the Invention

The present invention relates generally to the field of Radio FrequencyIdentification (hereinafter, “RFID”) systems, and more particularly toadvanced RFID systems which employ battery supported RFID tags allowinga higher degree of active behavior in tags.

B. Background of the Invention

The applications and importance of RFID technology has significantlygrown in recent years due to a number of reasons including improvementsin IC processes, RFID standards development, government allocation ofincreased spectrum for RFID, and growing awareness of the value ofautomated tracking of assets. During this growth, RFID systems haveprogressed from relatively simple, lower-frequency systems to includemore complex systems that operate in the longer-range Ultra-HighFrequency spectrum. The lower-frequency, generally inductively-coupledsystems are usually referred to as Low Frequency (hereinafter “LF”,approximately 100-150 KHz) and High Frequency (hereinafter “HF”,typically 13.56 MHz) systems. These system generally operate from rangesof a few centimeters to approximately 1 meter, and are limited in rangedue to the physics of “near field” communications that do not rely on apropagating electromagnetic wave. The systems operating in theUltra-High Frequency (hereinafter “UHF”, typically 800 to 1000 MHz)range can have longer ranges due to more favorable physical propagation.

Passive LF, HF, and UHF RFID systems comprise tags that operate withoutbatteries and effectively leverage power that is wirelessly receivedfrom an RFID reader to communicate information back to the reader. Inthe UHF case, this process is typically called “backscatter” and allowsa passive tag to communicate with an RFID reader over limited distances.Because these tags are effectively powered by the field of an RFIDreader, the distance the tag can communicate is limited by its own powerconsumption. As a result, passive UHF systems generally operate withpractical ranges of several meters.

UHF passive backscatter RFID has enjoyed success in recent years due tothe availability of small geometry integrated circuit processes thatenable low cost integrated chips to go in the tags, low cost tagproduction and test processes, greater market awareness of RFIDbenefits, and effective standardization. In the standards arena, theEPCglobal™ Gen 2 standard has been particularly successful in growingthe supply chain RFID market. The International Organization forStandardization (ISO) has also recently moved to adopt the EPCglobal™Gen 2 version 1.2.0 passive standard and to extend it with higherperforming battery assisted RFID under its ISO/IEC 18000-6C standard.

The EPCglobal™ Gen 2 standard is a “reader talks first” architecturewhere the reader “selects” categories of tags with a “Select” command.The chosen tags then are entered into an “interrogation round” or “queryround” with a “Query” command. The Query command provides the tags witha random number size range indictor, and the tags generate a randomnumber within that range that is then assigned to a counter to create a“time slot” for each tags. Tags seldom select the same time slot whenthe random number range is large compared to the number of tags withinrange of the reader. In the query round a “QueryRep” command theninstructs each tag to decrement its random counter. When the tag countreaches zero, the tag “replies” with a 16 bit random number. If thereare no collisions between tags that happened to have the same time slot,the reader replies with the same random number and the associated tagthen transmits its unique identifying code. The process of separatingtag replies in time effectively allows the tags to use low costreceivers that are on the same RF “channel” as far as the relativelybroadband tags can tell. The process of the reader establishing contactwith a particular tag is called “singulation”, and the process of thereader further interacting with the tag is called “access” In access,the reader can read and write data to tag memory.

To allow multiple readers to be in communications with overlappingpopulations of tags, EPCglobal™ Gen 2 based standards use the concept of“sessions”. A two bit variable called the “session” or “session code”identifies 4 sessions, each of which have a “Session” or “Inventory”flag with a “session state”. The flag is normally described as havingstate symbolic logic state “A” or “B”, where each of these can be mappedin a particular tag's internal design to either Boolean state “0” or“1”. Either “A” or “B” can be used to indicate that a tag has beensuccessfully singulated or “inventoried”, though it is more common inpractice to use “A” for “not yet inventoried” and “B” for “recentlyinventoried”. When up to four readers are simultaneously in range of atag, these readers may each singulate the tag in near real time (the tagcan be in the singulation process with all) via each reader using adifferent session, that is, using a different session code and the flagassociated with that session.

Despite UHF RFID systems having extended range as compared to LF and HF,there are many applications needing a still longer operating range whilealso maintaining high reliability. Also, the limited range of thepassive tags when the tags are in motion leads to limited time toconduct operations such as memory reads and writes. Sometimes even thetime to send separate Select and Query commands for selective tag accessis not reliably available.

Active RFID systems extend range by providing a power source and fullfeatured radio on the tag. “Full featured” is intended to mean a highlysensitive and selective (interference rejecting) receiver and activetransmitter whereby the tag creates its own transmit signal. Theseactive systems can achieve ranges of hundreds of meters, but costsignificantly more than passive systems. Additionally, the operationallife of the active systems is limited by the batteries deployed withinthe tags and the ability to replace these batteries over the life of thesystem. Some applications, such as tracking of military supplies, canabsorb the relatively higher cost of these active systems, but manyothers cannot.

To provide an intermediate level of performance between fully passiveand fully active RFID systems, there has been over the last few years amovement to introduce “battery-assisted” or “semi-passive” RFID systems.These systems utilize the UHF band and extend upon passive tags byproviding tag operating power from a compact battery such as a coincell, thus enhancing range by eliminating the requirement for the tag toreceive sufficient RF signal power to actually power itself from thesignal. The tag may also utilize baseband signal gain to further enhancesensitivity. The tag maintains the use of a simple and low power“backscatter” transmitter that operates by modulating a reflection of areader provided RF signal back to the reader. Standardization effortshave been underway within the International Standards Organization (ISO)to add semi-passive RFID technology to its EPCglobal™ Gen 2 based UHFRFID standard, ISO/IEC 18000-6C. The applicant is an active member ofthis organization and has contributed significantly to this particulareffort.

1. DEFINITIONS

For the purposes of this invention, the following RFID tag types aredefined by class. The RFID tag descriptions refer to UHF RFID tagsgenerally operating in industrial, scientific, and medical bands withother short range radio applications, or in specialized RFID bands from400 to 1000 MHz (most commonly 800 to 1000 MHz).

1. Passive or Class 1. In these systems, tags operate without a batteryand are powered by an incoming reader field of a reader. A tag has adetector which converts RF energy into DC energy to power associatedintegrated circuitry within the tag. Tag sensitivity is generally on theorder of about −5 dBm to −20 dBm, and reader sensitivity is on the orderof about −60 to −80 dBm. Practical ranges are generally 1 to 5 meters.The system is generally “forward-link limited” due to the modestsensitivity of the tag.

2. Passive plus security or Class 2. These systems feature the sameradio link technology as Class 1, but with added memory and security,and sometimes other features such as sensors.

3. Semi-Passive or Class 3. These systems feature a small battery (e.g.,lithium manganese dioxide coin cell), for providing power to the tag,thus relieving the tag of very close proximity requirements to thereader. The tag receiver will generally still be wide-band detectorbased, though optionally improved by the use of active gain, and the tagtransmitter will still use backscatter modulation. A well designedSemi-Passive tag may have tag sensitivity of up to approximately −60 dBmwithout an RF amplifier. A well engineered Semi-Passive system can havefree space range of several hundred meters and practical ranges ofseveral tens of meters. However, due to asymmetric backscatter linkphysics that favors the forward-link from reader to tag, these systemswill typically be “reverse-link” limited by the sensitivity of thereader receiver. The system may also be limited by interference seen ateither the tag or the reader.

4. Semi-Active or Class 3 Plus. These systems supply an optional activetransmitter in the tag to substitute for backscatter transmission. Thisrelieves the reverse-link limit of the Class 3 link, and with theaddition of an RF amplifier in the tag creating tag sensitivity in therange of −70 to −80 dBm (U.S. bandwidth) generally results in anapproximately “balanced link” where approximately the same link loss isallowed in both directions. For example, a link employing a readertransmitting a maximum effective radiated power of +36 dBm (the currentlimit for U.S. operation) and a tag sensitivity of −75 dBm can allow upto 111 dB of total link loss in the forward link. If the readersensitivity is −110 dBm (achievable when the carrier does not have tomaintain a carrier due to the transmitter providing its owntransmitter), and the tag transmits 0 dBm, then the reverse link losscan be up to −110 dB. Class 3 Plus systems are not currently fielded,but they are the only class that has almost near perfect matchingbetween forward and reverse link performances, and there are compellingtechnical and economic reasons to develop them.

5. Fully-Active, Active, or Class 4. These systems use fully functioningradios at the tag with receiver bandwidths similar to spectraloccupancies of reader transmit signals, thereby allowing highersensitivity and interference rejection at the tag. They also use tagtransmit carriers generated on the tag that do not have to decline intransmit power as range increases, which is an inherent weakness ofbackscatter systems. These systems currently exist and function well,although the tags are approximately an order of magnitude higher in costthan semi-passive systems, and about two orders of magnitude higher incosts than passive systems. An enhancement to the state of the artpresented in this disclosure is the part time use of Fully-Active radiocircuitry in the tag in combination with high performance Semi-Passivecircuitry that is used under most operating conditions, thus maximizingbattery life while providing additional performance when needed.

6. Battery Assisted Passive tag, or BAP tag. This term specificallymeans a battery assisted tag that maintains a backscatter transmitter,or a Class 3 tag.

7. Battery Assisted Tag, or BAT. This term also commonly refers to a tagwith battery assisted tag receiver enhancement, while still maintaininga backscatter based tag transmitter. The term was originally coined tospecifically refer to Class 3 operation and to distinctly mean nothaving active radio features on the tag. However, it is envisioned herethat Class 3 will become a battery saving “base mode” for Class 3 Plusand Class 4 tags that use Class 3 when the link is sufficient, andprogress to the active modes as needed. Thus, the use of the term “BAT”may in the future come to refer to any tag with battery assisted tagreceiver enhancement. In this disclosure a BAT may thus refer to a Class3 Plus or Class 4 tag that supports Class 3 operation, with the optionof using the more advanced Class 3 Plus or Class 4 modes when linkconditions require that higher performance.

8. Hibernation or Hibernate Mode. A state of low power consumption(sleep) in which a tag can listen for an “activation” command to awakenit to “normal mode” for full communication and operation. Class 3, 3Plus, 4 and other tags may optionally implement a hibernate mode.

8. Power Leveling. A wireless industry term applied to generalintelligent control of transmitter RF power levels. Transmit powercontrol is a commonly used means of controlling interference in densewireless system such as cellular telephony.

SUMMARY OF THE INVENTION

Systems, methods and devices are described for battery supported RFIDsystems that can both provide an intermediate level of performancebetween passive and Fully-Active systems, which provide for extension toactive operation, and which for provide for reliable interferenceresistant operation in the presence of passive, battery assistedpassive, and Fully-Active tags. This allows for flexibility acrossvarious system implementations, which enables an RFID infrastructure toefficiently operate with tags covering a wide range of possibilities,and where users can select the tag performance corresponding to aparticular application.

Simply adding tag sensitivity and greater range to the existingstandardized passive RFID system design results in escalatingdifficulties. These include the fact that only four session codes arehighly interference prone in that it is insufficient to fully separatethe readers that may be in range of a sensitive tag. Also, the statemachine governing passive tag operation is interference prone in that itwas not designed to allow for rejecting reader commands from distantreaders that more sensitive tags can hear. Greater tag sensitivities,and the improved reader sensitivities that are needed to operate withmore sensitive tags, are also subject to interference both within andfrom outside the RFID system.

Certain embodiments of the present invention provide improved batteryassisted RFID systems and tag designs by facilitating better levels ofsensitivity than prior art tags, optionally using a plurality of dynamicrange states to cover the necessary tag receiver dynamic range. Theseembodiments support methods of tag receiver circuit training to protectthis tag sensitivity, allow selection of the proper tag receiver dynamicrange state, and also allow receiver AC coupling training and symbol andframe synchronization.

Command sets and tag state machine designs are provided to allowimproved interference control to protect the improved sensitivity ofreaders (interrogators) and tags, and to allow convenient extension ofactivation and other commands for incorporation of more active behaviorsin the tag. This allows for a scaleable system design for variousperformance and price tags whereby users can select the tag performanceneeded under a common system and infrastructure design. In certainembodiments, additional interference control may be achieved throughRFID specialized methods of reader and tag transmit power level control.

A preferred tag sensitivity at ultra-low tag power consumption isachieved using a transistor detector based “square law” receiver modethat improves the possible sensitivity of a “direct detector” receiverarchitecture. The term “direct detector” is intended to mean a detectorthat converts input RF power to baseband voltage or current withoutneeding a power consuming local oscillator. A “square law” receiver, asopposed to a peak detector direct detector receiver, is a directdetector architecture receiver where output current or voltage isproportional to input current or voltage squared.

In various embodiments of the invention, a biased transistor-basedsquare law detector is employed within the BAT receiver that efficientlyharvests current from an RF signal by reducing the effects of voltage orcurrent division experienced in prior art diode-based square lawdetectors. The power output of a square law receiver is proportional tothe square of the RF power input. This inherently tends to limit thedynamic range achieved by the square law receiver. In prior artdiode-based square law receivers, their more limited sensitivity hastended to mask this dynamic range limitation. However, improvements inRF sensitivity according to various embodiments of the invention resultin the need to improve baseband dynamic range. Accordingly, certainembodiments of the square law receiver may be especially suitable tooperate within a system employing several dynamic range states with atotal larger dynamic range. This may ensure that full sensitivity neededfor longer range and lower reader transmit powers used to limitinterference are simultaneously available with the higher tag receivepowers that occur in very close range operation. This allows the systemto achieve the practical requirement of operating over a wide range ofdistances from a few centimeters to tens or even hundreds of meters.Also disclosed are protocol features allowing rapid selection of theproper dynamic range state for conducting a particular communications.

Certain embodiments of the present invention provide “tag training” andframe synchronization methods that may be applied to battery assistedRFID tags covering large dynamic range. For example, protocols thatallow selection of optimum preferred dynamic range state are disclosed.These methods also allow for improved AC coupling training, orequivalently, for settling of a slicer (comparator) referenceacquisition system that provides a slicing comparator reference levelthat adapts to the signal strength variation of the RF input. Use ofwell-controlled comparator reference levels may be particularly relevantto a preferred sensitivity, and are provided for by various disclosedtraining methods. These training methods are extended herein to alsoprovide for reliable identification of tag “wake-up” or Activationcommands by use of pseudo-random or “PN” data sequences. The PN sequencemethod may also be used to provide for frame synchronization flags usingjust the zero and one digital symbol alphabet; as opposed to prior artmethods using additional specialized symbols with different length andspectral content from the standard symbol set. Prior art methods usinglonger symbols as flags require AC coupling methods that have lower highpass corners, which are more expensive of integrated circuit die areaand take longer to train.

Embodiments of the present invention also provide a command structurethat allows for improved operation using battery assisted tags. Anexample of such an improvement is “tag-to-reader locking” where the tagresponds only to the reader which awakened or “activated” it from a lowpower hibernate mode. This mode effectively overcomes the limitation ofonly four sessions by introducing a replacement that can in a preferredembodiment provide up to 256 “effective” sessions while remainingbackward compatible. Various embodiments also provide for anintermediate level of interference rejection by introducing “sessionlocking” whereby the RFID system makes improved use of the interferenceresistance that is possible with only four sessions. For example, undercurrent EPCglobal™ based standards, the Select command will pull a tagout of most logical states in its state machine and send it back to thestarting point “Ready” state. In a preferred embodiment, the presentinvention introduces “Session Locking” whereby during the activationprocess a deliberate decision is made as to whether to allow suchinterference by Select commands, or whether to reject it with an“Interference Resistance” or “Session Locking” mode. This decision isenforced via a flag state introduced into the Activation command. Asuggested embodiment was an explicit “Interference Resist Flag” in theActivation command. The method subsequently chosen for standardimplementation was to make double use of another suggested flag whichplaced a “care/don't care” status for checking the state of the sessionflag as reused in hibernate mode and controlled by a newly introducedpreferred embodiment precision timer. This timer controlled the state ofthe session flag in hibernate mode, thus reusing this flag that innormal mode indicated inventoried state. The standardized compromise wasto also interpret a “care” on the inventory flag state as alsocommanding “Session Locking” to be in effect following activation.

Though tag-to-reader locking has so far been drafted into standardizedform only for the case of the tag-to-reader locking to apply while innormal mode following activation, it may also advantageously be appliedin hibernate mode. In that embodiment, the hibernating tag keeps trackof the last reader ID code that awakened it, and only allowsreprogramming by a reader providing that same code within an Activationcommand. An additional flag in the Activation command may be used forcommanding hibernation mode tag-to-reader locking, or a single suchcontrol flag could command tag-to-reader locking to apply in both thehibernate and normal modes.

New Query commands and Query command behavior are also introduced whichprovide expanded functionality, via providing a built in “mini-select”functionality for quickly identifying groups of tags with features incommon for fast access. This eliminates the need for slower use of the“Select” command to screen tag populations for inclusion in Queryrounds. It is backward compatible to future passive RFID systems. Thecommand set improvements disclosed may also provide scalabilityfunctionality that allows the command infrastructure to accommodate moreadvanced and later-deployed devices within the RFID system, such asthose featuring higher degrees of active behavior.

Embodiments of the present invention further provide battery managementtechniques in which a tag is duty cycled to reduce current drain fromthe battery. Such techniques may use either or both of duty cyclecontrol fields in commands, or files on the tag whereby duty cycle isspecified. Such files may be altered by standard memory access commands.Other functionality, such as interference limiting transmit powercontrol or “power-leveling”, is also supported by certain commands andfeatures.

In various embodiments of the invention, tags can be “awakened” fromhibernate to a higher power operational or “normal” mode by an ActivateCommand. Activation command structures and feature sets are disclosedthat extend the previous state of the art with new capabilities. Oneexample is the previously mentioned method of optimizing the channelbandwidth to that only needed for the data symbol set in use viaelimination of special and rarely used symbols, such as unusual framesynchronization flag symbols with frequency content significantlydifferent from the standard symbol alphabet. This is achieved by use ofPN sequences for activation identification and frame synchronization,and use of these PN sequences in ways not used in the prior art. Anotherexample is to specify flags and flag states to be used in power levelcontrol operations. Another example is use of programmable tagsensitivity to raise it above the particular noise environment.

Various embodiments of the invention expand the Activation command toscale with future deployments by having extendable fields that are ableto convey information potentially needed by future devices. For example,those fields may contain the class or classes of devices awakened by theactivation code, in particular tags with active functionality. Such tagsmay need additional control fields in activation, such as activetransmit frequency, active transmit power, active receive frequency,active tag receive sensitivity, and active receiver duty cycle.

In certain embodiments of the invention, the battery assistedbackscatter tag system is shown to be “reverse link limited,” and foroperating conditions where this limit is unacceptable, the system mayswitch to active transmit at the tag. A preferred square law receivermay be supplemented by an RF Low Noise Amplifier to establish a“balanced link,” where both forward and reverse links have path lossesthat are approximately equal. For example, an active transmit powerlevel on the tag may be supplemented to enable a balanced link byactivating a tag battery(s), such as lithium manganese dioxide coincells. Thus, a preferred square law receiver in conjunction withpart-time active transmitter provides a blend of maximum battery lifeand high performance.

The battery supported backscatter system with a square law tag receivermay lead to a reverse link limited situation where full power readertransmission would allow for longer reader-to-tag range thantag-to-reader range. The reason for this asymmetric link physics ofbackscatter RFID systems is the decreasing tag transmit power withincreasing range from the reader. In various embodiments of theinvention, this physical fact is used to control interference by usingvariable power levels on the forward and reverse links. Generally, morepower is used to support the backscatter reverse link than is used forthe forward reader to tag link. Since it is the forward transmissionsthat cause the most interference, this step limits interference.

In various embodiments of the invention relative to Class 3 powercontrol, an Activation command and other commands are provided in whichfields indicate the reader transmit power used in the forward link andthe difference between the power used in the forward link and thebackscatter supporting reverse link. This information may becommunicated by separate fields, or by a single field in which a bitspecifies where the particular information is forward reader power orthe difference between forward reader power and reverse reader power.Alternatively, separate fields may communicate reader forward power andreader reverse (backscatter supporting carrier) power. These Class 3power control methods also apply to Class 3 Plus and Class 4 tags withoptional active behavior when those tags are operating in their Class 3backscatter fall back mode of operation.

The information on the difference between forward and reverse readerpower allows a BAT to adjust the percentage of incident power reflectedas backscatter power in order to meet regulatory requirements and reduceinterference, which may be relevant in various environments including aClass 3 mode. The information about forward reader transmit power allowsthe tag to calculate path loss and determine an appropriate tag transmitpower to use during an active mode, such as in Class 3 Plus (square lawreceiver plus part-time active transmitter) or Class 4 (part-time fullyactive transmit and receive) modes. In these active modes there is noreader carrier to backscatter as the tag creates its own transmitcarrier.

Embodiments of the present invention also employ power leveling methodsthat optimize the amount of transmitted power from a reader in relationto that needed to reliably support the desired communications. Thisimprovement of transmitted power allows the reader to potentially reducethe amount of interference caused to other RFID readers and other radiosystems, and reduce RF energy exposure to individuals or materialproximate to the reader.

Embodiments of the present invention also provide for discovery andaccess of tags using a plurality of “mini-rounds” (tag accessed in setsas a function of approximate similar range) and optimum link operationbased on the unique physics of backscatter RFID, which allows forsequential discovery and access of tags based on their relative distancefrom a reader. Within each mini-round, the above-discussed increase ofbackscatter supporting reader carrier power as compared to forward linkpower is used to resist the effects of the asymmetric link physics. Tagstate controls may be employed within this discovery process such thatdiscovered tags are placed in a non-responsive state which prevents themfrom responding to a reader in subsequent rounds of higher power andlonger range discovery. The time period in which such tags do notrespond is controlled by accurate timers, which may be programmed by thereader to adapt to the actual situation.

The discovery and accessing of tags by a reader may involve the RFIDreaders first transmitting at low power, interrogating tags in a lowpower mini-round, typically using an increase in reader carrier powerduring backscatter to offset the asymmetric link physics, putting thesuccessfully accessed tags into a non-responsive state for a period oftime that is accurately controlled by a timer that has beenappropriately programmed for the actual environment, and then increasingthe power to conduct another mini-round at a higher power (i.e., longerdistance). This process may be iteratively repeated until the maximumrange, determined by the weakest link (usually the reverse link), hasbeen reached. Still higher ranges may be supported by using the Class 3Plus capability, if available, for more distant tags to link back to thereader. This process allows identification of tags and access of theirmemory and features over a wide range of distances without unnecessarilyusing higher power than necessary to access other tags that are closerto the reader, thus reducing interference.

A preferred embodiment is for the power leveling used in the activeClass 3 Plus and Class 4 modes to be fully integrated with the hibernatemode of Class 3 for maximum total system performance. Thus activationand normal mode command set designs are disclosed supporting methods ofpower level control that begin during the activation process andcontinue into normal operation.

Although the features and advantages of the invention are generallydescribed in this summary section and the following detailed descriptionsection in the context of embodiments, it shall be understood that thescope of the invention should not be limited to these particularembodiments. Many additional features and advantages will be apparent toone of ordinary skill in the art in view of the analyses, drawings,specification, and claims herein.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference will be made to embodiments of the invention, examples ofwhich may be illustrated in the accompanying figures. These figures areintended to be illustrative, not limiting. Although the invention isgenerally described in the context of these embodiments, it should beunderstood that it is not intended to limit the scope of the inventionto these particular embodiments.

FIG. 1 generally illustrates communication between an RFID reader and abattery assisted RFID tag according to various embodiments of theinvention.

FIG. 2 is an exemplary plot showing a relationship between readersensitivity needed to successfully maintain communications with theillustrated tag sensitivity according to various embodiments of theinvention.

FIG. 3 is an exemplary plot showing the reader carrier power needed inthe forward link minus the reader carrier power needed reverse link atthe same range in order to successfully support communications as afunction of reader-to-BAT range according to various embodiments of theinvention.

FIG. 4A is a circuit diagram illustrating a prior art biased diode-baseddetector.

FIGS. 4B and 4C are plots showing a square law zone and a lineardetection zone of the prior art diode-based detector of FIG. 4A.

FIG. 5 is a circuit diagram illustrating a small signal model for thediode-based detector.

FIGS. 6A-6E are circuit diagrams and relevant operating curves of animproved transistor-based detector using an NPN bipolar transistoraccording to various embodiments of the invention, and small signalmodels thereof used to analyze signal harvest.

FIG. 7 illustrates a transistor-based detection model capturing basebandsignal and noise according to various embodiments of the invention.

FIG. 8 is an exemplary plot showing bipolar transistor sensitivity (at aparticular data rate) as a function of the detector bias currentaccording to various embodiments of the invention.

FIGS. 9A and 9B illustrate exemplary multiple dynamic range states of adetector according to various embodiments of the invention.

FIGS. 10A through 10G illustrates symbol-based training and slicerreference acquisition according to various embodiments of the invention.

FIG. 11 illustrates possible continuous wave training according tovarious embodiments of the invention.

FIG. 12 is a chart showing frequency spray caused by symbol-basedtraining, which may be avoided by the CW based training of FIG. 11.

FIG. 13 presents an exemplary method to construct a hibernate modepreamble according to various embodiments of the invention.

FIG. 14 presents a different exemplary method to construct an examplenormal mode preamble that allows for dynamic range state selection andAC coupling training, and a complete PN sequence as a frame marker orsynchronization flag according to various embodiments of the invention.

FIG. 15 presents a cross-correlation of a preferred 15 bit PN sequenceframe synchronization flag with a preferred 21 training bit PN trainingsequence and the same 15 bit frame synchronization sequence according tovarious embodiments of the invention.

FIG. 16 shows a preferred method of bit destuffing according to variousembodiments of the invention.

FIG. 17 shows exemplary curves of False Command Response due to biterrors when the stuffed data stream suffers a bit error that causes anaccidental PN flag to be seen at the tag.

FIG. 18 shows curves of Battery Life Acceptability Factor (“BLAF”) thatindicate where PN sequence design acceptably achieves false wake uprates that dissipate less than 1% of tag battery life according tovarious embodiments of the invention.

FIG. 19 illustrates a Semi-Passive Class 3 Activation command structureaccording to various embodiments of the present invention.

FIG. 20 illustrates a Semi-Active or “Class 3 Plus” Activation commandstructure according to various embodiments of the present invention.

FIG. 21 illustrates an active or “Class 4” Activation command structureaccording to various embodiments of the present invention.

FIG. 22 illustrates an expanded reader Activation Control field that maybe integrated within an Activation command structure (such as thoseillustrated in FIGS. 19-21) according to various embodiment of thepresent invention.

FIG. 23 illustrates an expanded Reader Information field that may beintegrated within an Activation command structure (such as thoseillustrated in FIGS. 19-21) according to various embodiments of thepresent invention.

FIG. 24 illustrates an expanded Target Field that may be integrated withan Activation command structure (such as those illustrated in FIGS.19-21) according to various embodiments of the present invention.

FIG. 25 illustrates an expanded Active Tx Set Up Field that may beintegrated with an Activation command structure for a tag featuringactive transmission (Activation commands of FIGS. 20-21) according tovarious embodiments of the present invention.

FIG. 26 illustrates an expanded Active Rx Set Up Field that may beintegrated with an Activation command structure featuring fully activereception (such as shown in FIG. 21) according to various embodiments ofthe present invention.

FIG. 27 illustrates a battery tag QueryRep command structure accordingto various embodiments of the invention.

FIG. 28 shows the battery tag command Next with a field that sets thehibernate sensitivity range to be used by the tag after it returns tohibernate mode according to various embodiments of the invention.

FIG. 29 shows the battery tag command Deactivate_BAT, which can commandhibernate mode sensitivity according to various embodiments of theinvention.

FIG. 30 shows an exemplary Class 3 Activation command expanded from thecommand shown in FIG. 18 by addition of a reader power information fieldaccording to various embodiments of the invention.

FIG. 31 is an exemplary expansion of the reader power information fieldof FIG. 29 according to various embodiments of the invention.

FIG. 32 is an exemplary expansion of a reader power information fieldthat may be used in the case of Class 3 Plus and Class 4 tags accordingto various embodiments of the invention.

FIG. 33 shows an exemplary Flex Query command that allows faster accesswhile still maintaining the ability to selectively bring tags into Queryrounds based on their basic types and attributes according to variousembodiments of the invention. The types are selected in the Tag TypeSelect field.

FIG. 34 depicts certain details of the Tag Type Select Field of FIG. 33according to various embodiments of the invention.

FIG. 35 illustrates a system that allows for a wide range of tagcapabilities to be reported by the tag to the reader, and for the readerto in turn select operating modes through tag settings according tovarious embodiments of the invention.

FIG. 36A provides a general example of a Battery Capabilities Word(“BCW”) according to various embodiments of the invention.

FIG. 36B provides an example of a Battery Settings Word (“BSW”)according to various embodiments of the invention.

FIG. 37 conceptually illustrates mini-rounds used to discover andcommunicate with a plurality of tags according to various embodiments ofthe invention.

FIG. 38 is a flowchart illustrating a method for forward power controlaccording to various embodiments of the invention.

DETAILED DESCRIPTION OF THE INVENTION

In the following description, for purposes of explanation, specificdetails are set forth in order to provide an understanding of theinvention. It will be apparent, however, to one skilled in the art thatthe invention can be practiced without these details. One skilled in theart will recognize that embodiments of the present invention, describedbelow, may be performed in a variety of ways and using a variety ofmeans. Those skilled in the art will also recognize additionalmodifications, applications, and embodiments are within the scopethereof, as are additional fields in which the invention may provideutility. Accordingly, the embodiments described below are illustrativeof specific embodiments of the invention and are meant to avoidobscuring the invention.

Reference in the specification to “one embodiment” or “an embodiment”means that a particular feature, structure, characteristic, or functiondescribed in connection with the embodiment is included in at least oneembodiment of the invention. The appearance of the phrase “in oneembodiment,” “in an embodiment,” or the like in various places in thespecification are not necessarily all referring to the same embodiment.

A. Battery Assisted RFID System and Tag

Class 1 passive systems are typically of only modest tag sensitivity andare typically “forward link limited,” meaning the reader-to-tag linkfails at a shorter range than the tag-to-reader link. Using a battery inthe tag improves tag sensitivity; however, due to the characteristics ofthe tag square law receiver, AC coupling and the implementation ofmultiple dynamic range states may be required, which may be accountedfor in a protocol according to various embodiments of the invention. Inmany instances, it is difficult for the weak reverse link to keep upwith the now much more capable forward link. However, the reverse linkmay be strengthened by advanced reader designs, such as ultra-low phasenoise local oscillator and maximum transmit carrier to reader receiverisolation. Even so, with tag sidebands close to the carrier frequency,the reader will usually be phase noise limited as to sensitivity. Thisis improved by the use of tag backscatter “subcarriers,” such as theMiller modulation mode of ISO/IEC 18000-6C, in which higher frequencysubcarriers move tag backscatter sidebands “down the phase noise curve”and thus improve reader sensitivity.

Making intelligent use of this improved sensitivity at both ends of thelink allows for reducing reader-on-reader and reader-on-tag interferencewith lower forward modulated power than the pure carrier used to supportbackscatter. Additionally, power control may be adjusted for betterinterference control brought about by the higher sensitivities of bothlinks. Other interference control measures, such as optional splitbandplans and time coordination, may also be used as well as optional orfull time active transmission from the tag for links or applicationsneeding a better link than is possible with backscatter.

FIG. 1 illustrates the system operation of a battery supported RFID tag.A tag 110 wirelessly receives a forward link signal 106 from an RFIDreader 100. The forward link signal is modulated by the reader 100 anddemodulated by the tag 110. Tag front end RF filter 144 provides RFchannel selectivity in the tag's low power RFID receiver, which is alimitation on RFID system performance since the tag is subjected tointerference from all transmitters above its sensitivity level in the RFband. In certain embodiments of the invention, high sensitivity RF tagsemploy wide band filters that are subject to interference and RFIDinterference control measures are described to cope with suchinterference. In other embodiments the front end filter 144 is narrowerand its frequency is adjusted in accordance with the regulatory regionof operation or the presence of interference. For example, the readermay sense the presence of interference due to cellular transmissionsnear the frequency of RFID operation, and command the tag to alter itsfront end filter frequency range. Such settings may be convenientlycommanded through memory writes by the reader to the Settings Filedisclosed herein. If the tag 110 includes optional Active Receiver 132,then tag 110 has access to a narrow-band interference rejectingreceiver.

When it is the tag's turn to transmit, and the tag is going to use Class3 mode, the reader provides a pure carrier that the tag can reflectivelybackscatter as reverse link signal 108 having an associated power leveland containing information to be received by the RFID reader 100. Theuse of the backscatter transmitter saves the cost and power consumptionof having an active transmitter on the tag.

Various embodiments of the invention include devices and methods thatenable improved reception of received signal at the tag, provide forpreferred modal operation (e.g., active or square law mode) of the tag,and provide for interference control within the total RFID system.Square law tag receiver 130 provides a certain level of sensitivity inthe tag. If this sensitivity is too much for the interferenceenvironment, then sensitivity control 128 controlled by Activationcommand 118 is used to limit sensitivity. When better sensitivity orinterference rejection is needed, then the tag may be commanded toswitch to optional active receiver 132, which is supported by battery112.

Power measurement capability in both square law receiver 130 and activereceiver 132 allows the tag to be aware of receive signal power, andcombined with power control information in normal command 120 allowscontrol of the power output of both backscatter transmitter 122 andoptional part time active transmitter 124. To improve battery life, dutycycle control 134 may be applied to place the receiver in a hibernationor normal mode.

Clock generator 136 may be used in conjunction with single crystal 114to generate return data rate clock, return subcarriers, controllerclock, data logger clock, and input reference frequency for thefrequency synthesizer of active transmitter 124 and active receiver 132.In certain embodiments, crystal 114 is a low cost and low power tuningfork type from approximately 20 kHz to 100 kHz, such as the common32.768 kHz “watch crystal.” Sensor and data logger 138 expands thenormal identification function of RFID to allow for market desiredsensor operations such as temperature logging of goods in the coldchain, and is improved via the timing precision of crystal 114.

Tag controller 140 may be of digital state machine or firmwareprogrammed microcontroller form, or a combination of microcontrollerplus hardware support such as subcarrier generation and receive symbolsynchronization. Hibernate control 142 may be a low powermicrocontroller or a dedicated state machine. Hibernate controller 142may include pseudo-random “PN” sequence flag correlator and bitdestuffer as later described.

PN flag usage will be fully described later and is a method that allowsa standard {0, 1} symbol set to serve as activation validity signalingand frame synchronization. Use of only the {0, 1} symbol set ispreferred over prior art methods of special longer symbols because itallows a reduced channel bandwidth, a reduced coupling capacitor sizeand die area, and a reduction of on-die flicker noise that limits tagsensitivity. Flicker noise is a particular problem in the case of CMOSintegrated circuit implementation.

B. Backscatter RFID Link Physics

A tag may receive energy from an RFID reader field and use this energyto power itself, receive information from the reader, and backscatter aninformation-bearing signal back to the RFID reader. In the case of apassive tag, the tag sensitivity will typically be limited by itsoperating power requirements.

One skilled in the art will recognize that the Friis equation may beemployed in mathematically calculating BAT receive power from forwardlink 106 as a function of reader transmit power and range, anddetermining BAT transmit power used in reverse link 108.

Power available to backscatter is generally four times (i.e., 6 dB) theavailable receive power at tag antenna 104 because when the tag antennaload is shorted for maximum reflection, its total impedance is cut inhalf and its current doubles. As a result, the backscatter powerincreases by a factor of four minus a small reduction caused by lossesassociated with switching across the antenna

The Friis equation giving available receive power as function oftransmit power is given as Equation 1 in simplified form assumingpolarization alignment between transmit and receive antennas.

$\begin{matrix}{P_{rec} = {{P_{tran}\left( \frac{\lambda}{4\pi} \right)}^{2}\frac{G_{trans}G_{rev}}{R^{n}}}} & {{Equation}\mspace{14mu} 1}\end{matrix}$

In this equation, “R” is the range in meters, “n” is the link exponent(ideally 2.0 for free space), transmit and receive antenna gains are“G_(trans)” and “G_(rec),”, and wavelength is “λ”.

Using the Friis equation, the average tag transmit power in reverse link108 may be written as:

$\begin{matrix}{P_{tran\_ tag} = \frac{4d_{c}{{\mathbb{e}P}_{tran\_ reader}\left( \frac{\lambda}{4\pi} \right)}^{2}G_{reader}G_{tag}D}{R^{n}}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

In Equation 2, “e” is the efficiency associated with the tag switching,generally about 0.5 to 0.8, but sometimes deliberately lower in order tomeet specific regulatory requirements. The term d_(c) is the duty cycleassociated with the return modulation, generally about 50% for AmplitudeShift Keying (“ASK”). G_(reader) is the gain of the reader antenna andG_(tag) is the gain of the tag antenna. The term “D” is representativeof “degradation” in the link due to non-ideal factors such aspolarization misalignment, multi-path cancellation, and absorption ofpower by materials in the link.

Power at the reader receiver antenna 102 from the tag is obtained bysubstituting Equation 2 back into Equation 1.

$\begin{matrix}{P_{rec\_ reader} = \frac{4d_{c}{{\mathbb{e}P}_{tran\_ reader}\left( \frac{\lambda}{4\pi} \right)}^{4}G_{tran}^{2}G_{rec}^{2}D^{2}}{R^{2n}}} & {{Equation}\mspace{14mu} 3}\end{matrix}$

Here the link exponent has doubled from “n” to “2n” because the tagreceive/transmit power fades once with the link exponent from the RFIDreader 100 to the tag 110 and fades again from the tag 110 to the RFIDreader 100. In other words, if the forward link is the inverse square,then the reverse link is inverse 4^(th).

Using the Friis transmission equation with reader transmit power and theminimum necessary power at the tag receiver as sensitivity Stag, themaximum forward link range as limited by tag sensitivity may be derivedas shown below.

$\begin{matrix}{R_{max\_ tag} = \left\lbrack {\left( \frac{\lambda}{4\pi} \right)^{2}\frac{{DP}_{tran\_ reader}G_{reader}G_{tag}}{S_{tag}}} \right\rbrack^{\frac{1}{n}}} & {{Equation}\mspace{14mu} 4}\end{matrix}$Similarly, by substituting reader sensitivity S_(reader) for readerreceive power in Equation 3, the below expression for maximum readersensitivity limited reverse link range is obtained.

$\begin{matrix}{R_{max\_ reader} = \left\lbrack {\left( \frac{\lambda}{4\pi} \right)^{4}\frac{D^{2}4d_{c}{\mathbb{e}P}_{tran\_ reader}G_{reader}^{2}G_{tag}^{2}}{S_{reader}}} \right\rbrack^{\frac{1}{2n}}} & {{Equation}\mspace{14mu} 5}\end{matrix}$

These forward and reverse link ranges define the range limitation of anRFID system, which clearly depend on the strength of the field receivedby the tag and its ability to reflect this energy as transmission power.It turns out that the battery assisted tag receiver can achievesufficient sensitivity that the tag backscatter and reader sensitivityset the link performance limits—a reverse link limited situation.Accordingly, greater care is needed in Semi-Passive Class 3 systems thanin forward link limited Class 1 systems in order to minimize backscatterlosses. In other words, the term “e” in the equations above must be asclose to 1.0 as can practically be achieved.

C. Tag and Reader Sensitivity Relationship

The performance of the system depends on both the reverse and forwardlink ranges. Ideally these ranges should be about equal to maintain thelink; otherwise, if one link is inferior then the better link simplygoes to waste. If the ranges are set to be approximately equal to eachother, the RFID reader sensitivity and the RFID tag sensitivities asfunctions of each other may be found as below.

$\begin{matrix}{{S_{reader} = {\frac{4d_{c}{\mathbb{e}S}_{tag}^{2}}{P_{tran\_ reader}}\mspace{14mu}{and}}},} & {{Equation}\mspace{14mu} 6}\end{matrix}$

$\begin{matrix}{S_{tag} = \sqrt{\frac{P_{tran\_ reader}S_{reader}}{4d_{c}{\mathbb{e}}}}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

These sensitivities are relative to total receive power, which includescarrier, any subcarriers used, and information bearing sidebands. It iscommon practice in UHF RFID to use subcarriers in the tag to readerlink. This practice moves the information bearing sidebands out infrequency, where there is less carrier phase noise in the reader todegrade reader sensitivity. In the case of commonly used “MillerModulation,” such as specified in EPCglobal™ Gen 2 version 1.2.0 andISO/IEC 18000-6C standards, the subcarrier is modulated via phasemodulation, and the subcarrier is thus suppressed. This avoids wastingenergy in the subcarrier, so that the only penalty for using thesubcarrier is the 3 dB penalty of splitting from 2 standard AM sidebandsto 4 sidebands (one on each side of the two subcarriers above and belowthe carrier). This 3 dB penalty is usually far outweighed by the gain inreader sensitivity achieved by getting “down the phase noise curve”(farther from the carrier in frequency, where phase noise is less) byuse of the subcarrier.

As Equation 6 illustrates and as is shown in FIG. 2, each time animprovement is made in the forward link 106 of one dB in tagsensitivity, the reverse link 108 must improve by 2 dB of readersensitivity to maintain the same link margin. Accordingly, the physicallimitations of an RFID reader 100 will typically be reached before a BAT110 reaches its sensitivity limits (these BAT sensitivity limits will bederived later). The relationship between required reader and tagsensitivities as graphed in FIG. 2 is very important in Class 3 RFIDsystem design. As is seen in FIG. 2, a −40 dBm tag sensitivity requiresa reader sensitivity on the order of −110 dBm in order for the reader tosupply a balanced link.

This result is shown in a different form in FIG. 3, where the differencebetween required reader forward power and required reader reverse power302 is graphed using the high tag sensitivities disclosed herein againstreader-to-tag range 300. The plot in FIG. 3 has the assumptions offree-space propagation, reader antenna gain of 6 dB, tag sensitivity of−60 dBm, reader sensitivity of −90 dBm, fade margins of 10 dB in theforward link and 10 dB in the reverse link, BAT backscatter loss of 5dB, BAT antenna gain of −2 dB, and a frequency of operation of 915 MHz.At a reader-to-BAT distance of 7 meters, the reader transmit powerrequired in the forward link is 1000 times smaller (−30 dB) than thereader CW transmit needed to provide for tag backscatter in the reverselink. Accordingly, one skilled in the art will recognize the need toimplement power control in the forward link, and for reader power togenerally be higher in the reverse link.

A factor in determining sensitivity limits is that in the absence of anRF low noise amplifier, it is usually the baseband signal-to-noise ratio(hereinafter, “SNR”) that limits the tag sensitivity. This baseband SNRlimits the signal integrity because the conversion efficiency from an RFinput to the baseband output of even a highly efficient square lawdetector is still fairly low. The small harvested signal then has tocompete against baseband noise which can be fairly significant. Forexample, a −60 dBm RF input power will be converted to baseband withabout a 40 to 50 dB conversion loss relative to the total carrier plusthe sideband power. The resulting baseband signal power output of about−100 to −110 dBm must then compete with baseband noise. If theelectronic noise floor is about 8 dB above thermal noise, then with thethermal noise in a 50 KHz bandwidth about −127 dBm, the noise is about−119 dBm as compared to the signal power at about −100 to −110 dBm(SNR=9 to 19 dB). Since detected output power is dropping as the squareof declining input power (2 dB per dB), the tag may be expected to hitnoise limited sensitivity not far below −60 dBm. This is found to betrue via experimental verification.

A detector may be designed to overcome the baseband noise limit bygenerating sufficient front-end RF gain prior to the detector. When thishappens, the receiver sensitivity will improve until it becomes limitedby the noise in the RF bandwidth of the receiver. For example, thethermal noise power in a 50 MHz bandwidth appropriate to a U.S. RFID tagis about −97 dBm. If the front-end filter has about a 3 dB loss and thelow noise amplifier (hereinafter, “LNA”) has about a 3 dB noise figure,then the receiver generally has approximately a 6 dB noise figure. Thisnoise figure may be increased by a small amount to account for back-endnoise within the receiver, so a reasonable estimate would be 8 dB. Thisresults in a receiver input noise from the LNA of about −97 dBm+8 dB=−89dBm.

An AM modulation mode requires about a 3 dB loss due to half the energybeing in the carrier and not in the information bearing sideband.Accordingly, the SNR ratio required is approximately 15 dB (typical 12dB relative to information bearing sidebands, plus this additional 3 dBof penalty). However, this is the demodulated SNR requirement, and dueto the square law behavior only half of this in dB is required at RF.Using a low noise RF amplifier, the Manchester coded AM sensitivitywould be expected to be limited to about −81 dBm [i.e., −97 dBm (50 MHznoise floor)+8 dB (Rx noise figure)+8 dB (total RF SNR needed rounded tonearest dB)]. It may degrade several more decibels with practicallimitations in circuit design.

This sensitivity could be moderately improved for a European tag with anarrower surface acoustic wave (SAW) front end RF filter appropriate tothe narrower European bands. As the current European band is 865 to 868MHz, a front end filter with a 3 dB bandwidth of approximately 5 MHzwould be appropriate. This would improve LNA enhanced noise limitedsensitivity from about −81 dBm to −91 dBm. These sensitivitycalculations illustrate the physical limits of certain detector-basedsquare law receivers using RFID bandwidths. These limits areapproximately 20 to 40 dB inferior to active receiver sensitivities, butthey are as much as 80 dB superior to passive tag sensitivities.

D. Diode-Based Detection and its Limitations

FIG. 4A is an illustration of a prior art biased diode-based detector.This diode-based detector has improved sensitivity over a peak detectorpassive approach by biasing it to overcome the turn-on voltage of thediode. The detector diode 408 has a small steady state DC current flowthat is provided by supply 406 and a first series resistor 404 and aload resistor 410. The small DC current “biases up” the diode so that itis always on, which allows relatively small input RF signals 400 tocause peak detection on the load resistor 410 and peak holding capacitor412. The load resistor 410 provides a discharge function of the peakvoltage captured on the holding capacitor 412 to follow the envelope ofamplitude modulated RF input signal 400. Additionally, and as will belater described in detail, signals that are too low for peak detectioneven with a biased detector can still be detected using the square lawprocess to be described.

FIG. 4B is a graph generally illustrating two zones of operation for adiode-based detector. The graph illustrates the output voltage 426 onthe detector as a function of the RF input carrier voltage 424. A firstzone of operation, identified as a linear or peak detecting zone 420,exhibits a linear relationship between the RF input carrier voltage 424and the output voltage 426. This first zone 420 is defined as operatingwhen the RF input carrier voltage 424 is above a particular powerthreshold value, which in certain examples of the diode-based detectoris approximately 10 μW (−20 dBm).

A second zone, identified as a square law zone 418, exhibits anon-linear squared relationship between the RF input carrier voltage 424and the output voltage 426. As the input carrier voltage 424 (which isproportional to the square root of the RF input power) declines to lowerRF powers, the peak detecting operation of the diode-based detectorstarts to function differently once the power threshold is passed. Thisthreshold is typically at the point where the diode conduction cyclebecomes noticeably greater than 180 degrees in response to the decreasein applied RF signal. In other words, this occurs for signalssufficiently small that the diode does not “turn-on hard” on thepositive half of the RF swing. For sufficiently small signals, the diodeconduction cycle actually becomes 360 degrees.

The non-linear behavior of the diode means that for very small signals,there is a non-linear increase in current induced forward through thediode for the upper half of the RF signal and a non-linear decrease incurrent through the diode for the reverse half of the swing. Thisnon-linearity causes a small shift in the detected baseband voltageacross the load resistor 410 and the holding capacitor 412, even forvery small signals. In this detection of very weak signals, the detectedvoltage is proportional to the input RF power, and more specifically, isproportional to the square of the RF voltage. The detection of thesevery small RF signals is shown as occurring within the square law zone418. If the output voltage of the detector is graphed as a function ofinput RF power, then the graph shown in FIG. 4C results.

A square law receiver may be implemented within an RFID tag to detectweak RF signals of the general levels described above. The square lawreceiver harvests a baseband voltage or current from a detector that isproportional to the RF power input to the detector. In particular, theoutput power of the receiver is proportional to the square of thisvoltage or current, and thus the output baseband power is proportionalto the square of the input RF power. This is the source of the term“square law receiver”, and is also the source of the physical fact thatoutput baseband power changes 2 dB per input RF power change of 1 dB. Aweakness of the square law receiver is that this 2 dB per dBrelationship limits the dynamic range of the output of the tag receiver.

The following equations provide a mathematical model to analyze signalharvest within a square law receiver. The total current “i” in anon-linear detector driven by an RF or other time domain signal “V_(in)”may be described by a generic Taylor Series expansion of a non-linearfunction as:

$\begin{matrix}{i = {I_{bias} + {\frac{\mathbb{d}i}{\mathbb{d}v}V_{in}} + {\frac{1}{2}\frac{\mathbb{d}^{2}i}{\mathbb{d}v^{2}}V_{in}^{2}} + \ldots}} & {{Equation}\mspace{14mu} 8}\end{matrix}$

The derivatives in the above expression capture the nature of theparticular non-linear device, meaning a device where the current flowdoes not follow Ohm's law but instead follows a non-linear function ofapplied voltage such as the exponential function typical of diodes,bipolar transistors, and MOS transistors operated in weak inversion. Thesecond order term is always significant since it is the lowest ordernon-linear term that can perform RF to baseband frequency conversion,and for smaller signals it will give a larger “harvest” than the higherorder terms. For an RF signal of V_(s) Sin(ωt) the below trigonometricidentity will allow evaluation of the 2nd order term:

$\begin{matrix}{V_{in}^{2} = {\left( {V_{s}{{Sin}\left( {\omega\; t} \right)}} \right)^{2} = {V_{s}^{2}\left( {\frac{1}{2} - {\frac{1}{2}{\cos\left( {2\omega\; t} \right)}}} \right)}}} & {{Equation}\mspace{14mu} 9}\end{matrix}$

If the second harmonic term in this identity is neglected and asubstitution of the DC term into the second order term of the TaylorSeries is done, for the DC current induced by an RF input of carrierpeak voltage V_(s) is:

$\begin{matrix}{{\Delta\; I} = {{\frac{1}{2}\frac{\mathbb{d}^{2}i}{\mathbb{d}v^{2}}V_{s}^{2}\frac{1}{2}} = {{\frac{V_{s}^{2}}{4}\frac{\mathbb{d}^{2}i}{\mathbb{d}v^{2}}} = {\frac{P_{in}Z_{in}}{2}\frac{\mathbb{d}^{2}i}{\mathbb{d}v^{2}}}}}} & {{Equation}\mspace{14mu} 10}\end{matrix}$In the above equation, Z_(in) is the impedance environment on the inputto the detector and P_(in) is the available RF input power. In order toreceive a benefit of passive voltage gain, an up-transform is performedfrom the antenna to the detector input. The effectiveness of this gain,and the size of the harvested currents that depend upon it, depends onthe structures and methods used in this up-transform.

As illustrated below, the bias current through a diode-based detector isgiven as a function of input voltage V_(in) as:

$\begin{matrix}{I_{bias} = {i_{o}{\mathbb{e}}^{\frac{V_{in}}{V_{T}}}}} & {{Equation}\mspace{14mu} 11}\end{matrix}$where V_(T)=kT/q and k is Boltzman's constant, T is absolute temperatureand q is the charge of an electron. Differentiating twice with respectto V_(in) gives:

$\begin{matrix}{\frac{\mathbb{d}^{2}I_{bias}}{\mathbb{d}V_{in}^{2}} = \frac{I_{bias}}{V_{T}^{2}}} & {{Equation}\mspace{14mu} 12}\end{matrix}$Substituting this into the expression for harvested DC current above,the following equation is derived for the diode-based detector:

$\begin{matrix}{{\Delta\; I} = {{\frac{V_{s}^{2}}{4}\frac{\mathbb{d}^{2}i}{\mathbb{d}v^{2}}} = \frac{I_{bias}V_{s}^{2}}{4V_{T}^{2}}}} & {{Equation}\mspace{14mu} 13}\end{matrix}$

As will be discussed below, the diode-based detector is unable toprovide all of this harvested current to a load. FIG. 5 illustrates asmall signal model of the biased direct detector diode shown in FIG. 4A,which shows inefficient current detection because a portion of thedetected current is wasted in the diode and not provided in the desiredload. The problem is that while it is desired that all the current flowin the load resistor 504, that current is limited by the bias resistor506 in series with the load resistor 304, and the tendency of thedetected current to be trapped by the small signal impedance 502 of thediode itself.

The detected current in the load resistor 504 is given by currentdivision as:

$\begin{matrix}{{\Delta\; I_{load}} = {\Delta\;{I_{total}\left( \frac{R_{diode}}{R_{load} + R_{bias} + R_{diode}} \right)}}} & {{Equation}\mspace{14mu} 14}\end{matrix}$By the Maximum Power Transfer Theorem, maximum power will be deliveredto the load when:

$\begin{matrix}{R_{diode} = {R_{load} + R_{bias}}} & {{Equation}\mspace{14mu} 15}\end{matrix}$

The maximum power occurs when half the current is diverted into theload. This is the maximum power available from the “source” T ofI_(detected) in parallel with R_(diode). It is evident that currentdivision will noticeably limit the harvested current within thereceiver. In fact, if R_(diode) was infinite, then doubling the currentdelivered to R_(load) would actually increase harvested power by asignificant 6 dB from the ideal matched case, and more in the case ofless than perfect matching. This 6 dB or more loss of harvested powermay significantly reduce the performance of certain RFID systems.

E. Improved Transistor-Based Detector

The transistor-based detector disclosed herein may overcome theabove-described deficiencies in diode-based detectors. FIGS. 6A-6Dillustrates a transistor-based detector with improved sensitivityaccording to various embodiments of the invention. The detector is ableto maintain the square law function, described above, while providingcurrent gain, relatively higher input impedance that improvessensitivity, and reduction of the lost part of the detected current bythe current divider within the diode detector. It also providesconvenient mirroring and deliberate dynamic range extending compressionof the detected current.

FIG. 6A is a basic circuit diagram of a transistor-based detectoraccording to various embodiments of the invention. The detectortransistor is 610, and the other transistors are for signal processingof the detected signal. Though a bipolar transistor is depicted, similardetection processes and advantages apply to MOS transistors also.Ra_(inout) signal 600 may have most of its voltage applied directlyacross the base-emitter junction of detector transistor 610, as thereare no other impedances in series with the base-emitter junction. Thereis no requirement to power match (the detection is a voltage drivenprocess) and the base-emitter small signal impedance may be greater thansource impedance 602, thus allowing most of the available RF voltagefrom source 600 to be directly applied to the detector base-emitterjunction.

In various embodiments of the invention, impedance up-conversion circuit605 inserted from source 600 to the base of 610 can also apply passivevoltage gain from source 600 to supply still more voltage swing. Thisdelivers still more harvested current, since the detected current isproportional to the square of the voltage swing across the base-emitterjunction and not actually the power delivered to the junction. As aresult of these factors, more detector current results, and a muchimproved percentage of the detected baseband signal is harvested in thetransistor detector 610.

The current multiplying behavior of transistor 610 allows the detectedcurrent ΔI_(b) in total base current 606 to be multiplied by the currentgain and generate a new and much larger detected current, ΔI_(c) whichcan be passed through arbitrary impedances to generate larger voltageswings. These impedances may be as large as other circuit limits allowbecause there is no need to impedance match a load resistor value todiode small signal impedance as was the case for the diode detector.Alternatively, this circuit design freedom allows the new current,ΔI_(c), to be mirrored and gained before passing through an impedancethat converts the current into a voltage signal. For example, the finalimpedance may be a very high impedance active load for maximumsensitivity. The net result of these many combined improvements is thata much larger signal is harvested, and much greater sensitivity results.

FIG. 6B illustrates the transistor detector 610 in a large signal modelform (including both bias currents and detected currents) with anon-linear detecting diode 620. The detecting diode 620 is formed by thebase-emitter junction through which the sum of bias current and adetected current 618 flows. A current source 622 represents thecollector current multiplier of the base current, which results in thecurrent 618 being multiplied by the current gain “β_(t)” of thetransistor into total collector current 624. The term β_(t) is usedbecause the commonly used term “β” is also used in the detector diodeliterature to refer to detector diode gain from input RF power in wattsto output current. The term “β_(d)” will be used herein for diodedetector gain in Amperes/Watt.

FIG. 6C is a simplified small signal model of the transistor detector610 according to various embodiments of the invention. As shown, thedetected current ΔI_(b) 628 flowing through the base-emitter junction isnot subject to the current division in the diode-based detectors.Although the detected current 618 flows in the self-impedance 626 of thebase-emitter junction, there is little, if any, loss associated withthis action. Instead, this “current trap” is used to advantage byproviding a change in base current to be multiplied to a change incollector current without loss, which constitutes a major advantage ofthe detector transistor over the detector diode. The final detectorcurrent ΔI_(c) 628 is now much larger than detected diode current wouldbe, and has the very large advantage of circuit freedom from divisionand matching. It may be mirrored and passed through arbitrarily largeimpedances to maximize sensitivity.

As previously discussed, the detected base current 628 may be multipliedby the current gain or β_(t) of the transistor, which is typically onthe order of 100 times. A potential problem may appear to be the normallarge variation in β_(t) of the transistor. But, if a bipolar transistoris biased at a nearly constant (except for variation caused by detectedcurrent) collector current, then transistor β_(t) and its undesiredvariation drop out as follows:

$\begin{matrix}{{\Delta\; I_{c}} = {{\frac{I_{base}V_{s}^{2}}{4V_{T}^{2}}{Beta}} = {{\frac{\frac{I_{c}}{Beta}V_{s}^{2}}{4V_{T}^{2}}{Beta}} = \frac{I_{c}V_{s}^{2}}{4V_{T}^{2}}}}} & {{Equation}\mspace{14mu} 16}\end{matrix}$

In this equation, V_(s) is the peak voltage of the sinewave RF signalV_(in). As shown by Equation 16, detected final current is not actuallya function of β_(t), which is a very desirable behavior since β_(t) isnot a very well controlled parameter. The detected current isproportional to total collector current just as it is proportional tototal diode current in the detector diode case.

To summarize, the transistor-based detector, as illustrated in FIG. 6,turns the disadvantage of harvested current dividing into the detector'sown impedance into an advantage by dividing all the current into thebase-emitter, and then harvesting it through the current multiplyingeffect from base to collector. The high input impedance of thetransistor also allows high “voltage gain” from RF input to transistorinput, further improving sensitivity. As previously mentioned, thecurrent can be mirrored, gained, and passed through high impedanceactive loads to attain sufficient voltage swing to drive a comparator.For example, a detected collector current of only 1 nA may be used todevelop adequate voltage to trigger a comparator using an “active load”impedance of 1 MΩ:

$\begin{matrix}{V_{out} = {{\Delta\;{IZ}} = {{\left( {1\mspace{14mu}{nA}} \right)\left( {1\mspace{14mu} M\;\Omega} \right)} = {1\mspace{14mu}{mV}}}}} & {{Equation}\mspace{14mu} 17}\end{matrix}$

Referring now to FIG. 6A, the detected current ΔI_(c) flows throughcompression resistor (explained below) into first mirroring transistor612. This transistor is “diode connected” as is known in the art, andits base-emitter voltage is also provided to second mirror transistorand output stage drive 632. Since transistor 632 has the same V_(be) as612, its collector current will almost exactly equal that in transistor612, thus generating a “current mirror”. It is also possible throughtransistor geometry adjustment or provision of resistance in theemitters of the two mirrored transistors 612 and 632 to provide “gain”in the mirror which may be less than one, equal to one, or greater thanone. A current mirror may thus serve an amplification function orattenuation function, either of which can be referred to as gain. When acurrent mirror drives a load or active load it may also be referred toas an “amplifier”.

The “active load” is the small signal output impedance of a transistor,as indicated in FIG. 6D Z_(out) 654. Because a bipolar transistor in theso-called “active region” as an almost constant current as a function ofV_(CE), it has a very high output impedance given by:

$\begin{matrix}{Z_{out} = \frac{\partial V_{CE}}{\partial I_{C}}} & {{Equation}\mspace{14mu} 18}\end{matrix}$

A similar very high output impedance applies to the “pinch off” regionof an MOS transistor. In either case active loads can provide muchhigher impedance at a given bias current than can a resistor with thesame voltage across it and current through it. Active loads are commonpractice in the integrated circuit design art, though not generally soin use with RFID detectors.

While the active load can provide for very high output impedance, itrequires special steps to control. When two transistor collectors (ordrains in the case of MOS transistors) are driving against each other,the currents in the two transistors cannot easily be perfectly matched.Thus, the output voltage of the common connection tends to go all theway high (almost to the positive supply) or low (almost to ground),depending on which current source provides more current. This is dealtwith in IC design by use of negative feedback that adapts one currentsource to almost perfectly match the other, so that the output voltagecan be at a desired level between the positive and negative (or ground)supply voltages.

To provide negative feedback in the case when Z_(load) 634 is an activeload, a low power feedback system may sample output voltage 636 andadapt a control voltage of Z_(load) 634 to adjust its current such thatoutput voltage 636 is forced to have a particular value. In certainembodiments, differential analog integrator circuit 637 is implementedwith a low power operational amplifier with one input as V_(out) 636 andthe other as adjustable V_(ref) 635. The negative feedback and very highDC gain of this circuit will set V_(out) 636 to have a DC level almostidentical to V_(ref) 635, which can then almost perfectly match thereference input of a comparator that is “slicing” V_(out) 636 into a“squared up” logic level signal.

V_(ref) 635 may be trimmed to make this match between the DC level inVout 636 and the comparator's reference or “slicing” level as accurateas desired, and to also trim out operational amplifier and comparatoroffsets. If the loop bandwidth of the control system formed byintegrator 637 is deliberately controlled to a precise low value, thenseveral advantages result. First, the low bandwidth will set the DCcontent of V_(out) 636 as just described, but will not significantlydistort the AC signal part of V_(out) 636, as the spectral components ofthe demodulated signal will be above the bandwidth of this control loop.Next, the control system action will have the benefits of centering thevarying signal around the reference input of the slicing comparator, aslong as time is left in the protocol preamble for such centering tooccur within the response time of this loop. The ability of the controlloop to significantly rebias the current pulled by an active loadserving as Z_(load) 634 also significantly extends the dynamic range ofthe output stage formed by 632 and 634. It forms a type of automaticgain control or AGC, whereby the average current pulled by Z_(load) 634may perfectly match the average current of drive transistor 632. This isimportant since under strong RF drive the detected current can actuallybe not only a significant fraction of standing bias current, but evenlarger than the standing bias current. Finally, when the loop bandwidthis below the spectral content of the demodulated signal component of thecurrent delivered by mirror source 632, the open-loop bandwidth of anoperational amplifier in integrator 637 can be low, allowing suchoperational amplifier to be very low power.

Because the integrator controlled active load servoes out or removes DCvariations in output voltage V_(out) 636, it introduces effectively ACcoupling in the signal path. Thus, certain embodiments are suited tomodulation forms such as Manchester where the spectral content of thesignal approaches zero energy density at low frequency. To allow forvariable data rates, the loop bandwidth of the integrator 637 controlloop may be switched with data rate, thus allowing fastest settling timein the preambles of the protocol.

It is also possible to use a predominantly digital control systeminstead of the integrator control system 637 of FIG. 6A. This requires afine trim on the current control input of active Z_(load) 634. Sincewith even fine trimming there is drift in V_(out) 636 over time andtemperature variation, this digitally controlled voltage will need to beregularly updated. One advantage for using a digital control system isthat its output may be held constant over a communication interval,deliberately avoiding control system response to the received data, fordata types such as PIE that have DC content that varies over time. Inthe case of PIE, since the symbols have different pulse width, the DCcontent is not 50% and it also varies over time as a function of thezero and one bit densities in the data stream. Such variation causes“hunting” in a constantly closed loop analog control system, just as itdoes for AC coupling, which is harmful to the received Bit Error Rate(“BER”). The digital control system action may be deliberately “frozen”over a communications interval to avoid this hunting, which may beconsidered to allow a temporary state of DC coupling. Herein thistemporary DC coupling mode is referred to as “quasi-DC coupling”.

Quasi-DC coupling may also be implemented in an analog control loop thatcontrols the active load output voltage 636 by “opening the loop” in themanner of a sample and hold circuit in accordance with variousembodiments of the invention. In the case of integrator 637, the sampleand hold capacitor may advantageously be the main integrator capacitor.

One skilled in the art will recognize that in describing feedbackcontrol of active loads without reference to specifically analog ordigital control loops, that such control loops may be analog or digitalor combined with features of both, including temporary open loopoperation.

For simplicity, Z_(load) 634 may also be a resistive load. In that case,then to properly slice the bias current of transistors 610, 612, and 632may be trimmed for precise adjustment, or the comparator reference inputmay be similarly trimmed. In certain embodiments, Z_(load) 634 is aresistor in a less sensitive high dynamic range state, and an activeload in a more sensitive lower dynamic range state. A resistive loadalso has the advantage of allowing pure DC coupling. This is superior asregards distortion control to the effective AC coupling of theintegrator controlled active load for modulations such as Pulse IntervalEncoding (“PIE”), which do not have zero energy density per Hz asfrequency approaches zero. Thus, in a multiple dynamic range state RFIDtag receiver based on embodiments of the invention, a high dynamic rangestate may use a resistive load for Z_(load) 634 and for a bettersensitivity low dynamic range state may use an active load for Z_(load)634.

Certain embodiments of the invention may anticipate a high dynamic rangestate being useable for Manchester and PIE, while a more sensitive lowdynamic range state is Manchester only.

Radios usually have to operate over a wide dynamic range of inputsignals, with short term variation that can exceed 40 dB due to signalvariation and multipath fade, and absolute variation that can exceed 100dB due to the addition of path loss to short term fading. The square lawdetector is inherently hostile to wide dynamic range, since its outputpower is proportional to the square of input power. This may bepartially addressed through the use of two or more dynamic range states,but even then it may prove difficult to cover the entire dynamic range.There are also situations where the use of multiple dynamic range statesis not allowed for in the system protocol. Deliberate compression of thecurrent output of the detector as a function of input RF power may beprovided, as is depicted in FIG. 6E, in order to provide for the higherdynamic range that expansion upon passive RFID sensitivity levelsrequires.

According to various embodiments of the invention, deliberate detectorcompression is provided to make use of the saturation region of abipolar transistor detector. This region, where collector current is astrong function of collector to emitter voltage, is shown as the curveparts below V_(CEsat) 652 of the collector current curves of FIG. 6D.This region applies for V_(CE) less than a few tenths of a volt, andthough it actually moderately increases with increasing I_(C), it ishere shown as a constant for simplicity. At these low voltages and at agiven base current, the collector current is approximately linearlyproportional to V_(CE). There is an analogous zone of operation for anMOS transistor, though it is normally referred to as the “triode” or“linear” region, where for a few tenths of a volt the drain current fora given gate to source voltage is approximately proportional to drain tosource voltage. For both bipolar and MOS transistors, at these lowvoltages the devices acts like a controlled resistor.

It is thus possible to introduce controlled collector to emitter ordrain to source voltage in the square law transistor detector such thatat low RF power levels the device is in its “active” (bipolar) or“pinch-off” (MOS) zone where the device has a relatively constantcollector or drain current with V_(CE) or Y_(DS), and where the detectorhas high conversion gain from RF input power to output detected basebandcurrent. But, as RF input power increases and detected current becomesan appreciable fraction of standing bias current, the collector toemitter or drain to source voltage may drop to push the transistor intoits saturation (bipolar) or triode (MOS) region where its detectedcurrent is smaller than it would be in the active (bipolar) or pinch-off(MOS) regions. A preferred embodiment for performing this function is tointroduce resistance in series with the collector or drain of thedetector transistor, as in shown with Z_(compress) 611 in FIG. 6A.Z_(compress) 611 forces the total current in the detector collector ordrain to follow a “load line” 648 as shown in FIG. 6D.

Various designs of this compression circuit call for first allowing azone of operation where the compression does not apply, meaning that thebias value of V_(CE) is safely greater than V_(CEsat). Then, a largeenough increase in I_(C) due to RF inducing ΔI_(C) increases the dropacross the compression resistor and begins pushing the transistor intosaturation.

For example, in FIG. 6D an increase from zero RF input power atoperating point A is provided to shift total collector current fromlower (bias) current curve 652 to higher current curve 644 (I_(BIAS)652+ΔI_(C1active) 654). This shifts the operating point from point A topoint B as shown in both FIG. 6D and FIG. 6E. In various embodiments,point B is designed to be a critical dividing line between thetransistor detector operating in its active region and in its saturationregions. Thus, in FIG. 6E there is a linear increase in total collectorcurrent as RF input power increase moves the operating point from pointA to point B.

As RF input power increases by a step above that associated with pointB, the increasing current through Z_(compress) 611 forces a drop inV_(CE) that takes the transistor from its active region to itssaturation region. This results in moving the operating point from pointB to point C in both FIG. 6D and FIG. 6E. There is a smaller increase(ΔI_(C2sat) 650) in total collector current than would occur(ΔI_(C2active) 646) if the transistor did not enter the saturationregion, and the difference between these amounts may be referred to asthe degree of “compression” that the circuit provides.

This behavior may be put on a first order analytic basis as follows. Theequation of the collector to emitter voltage of the load line 648 ofFIG. 6D as applied to detector transistor 610 of FIG. 6A may be writtenas:

$\begin{matrix}{V_{CE} = {V_{CC} - V_{BE} - {I_{C}Z_{compress}}}} & {{Equation}\mspace{14mu} 19}\end{matrix}$

In Equation 19, V_(BE) is the base-emitter voltage of diode connectedmirror transistor 612 of FIG. 6A. I_(C) is the total collector current,whether in the active or saturated region. When this equation is appliedto the active region, current variables may be subscripted as “active”and when applied when current in is the saturation region, then currentvariables may be subscripted as “sat”.

The collector-emitter voltage of the transistor is forced to obey thisequation as well as the transistor's operating equations. Though thetransistor operation is non-linear, for a first order calculation it maybe assumed that the current-voltage relationship of the transistor insaturation is approximated by:

$\begin{matrix}{V_{CE} = \frac{V_{{CE}\mspace{11mu}{sat}}I_{Csat}}{I_{C\mspace{11mu}{active}}}} & {{Equation}\mspace{14mu} 20}\end{matrix}$

In equation 21, V_(CEsat) is the dividing line between the active andsaturation zones, which is approximately 0.2 volts. I_(Csat) is avariable describing the actual current in saturation under a particularV_(CE)<V_(CEsat), and a particular base current. I_(cactive) is thetotal collector current when V_(CE) is greater than or equal toV_(CEsat). If Equation 19 is set equal to Equation 20, then thecompressed total current in saturation is found to be:

$\begin{matrix}{I_{C\mspace{11mu}{sat}} = \frac{V_{CC} - V_{BE}}{\frac{V_{{CE}\mspace{11mu}{sat}}}{I_{C\mspace{11mu}{active}}} + Z_{compress}}} & {{Equation}\mspace{14mu} 21}\end{matrix}$

When Equation 21 is applied to operating point C of FIGS. 6D and 6E,then it is written as:

$\begin{matrix}{I_{C\; 3{sat}} = \frac{V_{CC} - V_{BE}}{\frac{V_{{CE}\mspace{11mu}{sat}}}{I_{C\; 3{active}}} + Z_{compress}}} & {{Equation}\mspace{14mu} 22}\end{matrix}$

In Equation 22, V_(CEsat) is assumed as constant (approximately 0.2 to0.3 volts), and I_(C3) active is the sum of the bias current and whatwould be the detected current at point C if the transistor was allowedto remain in the active region. Equation 22 defines the curve of FIG. 6Egiving total detector output current for RF input powers greater thanP_(in) 678 associated with point B.

In order to use these small voltages and attain enhanced sensitivity,fully trained AC-coupling and low offset or trimmed comparators may beimplemented according to various embodiments of the invention. A naturalconsequence of operation within square law mode is that multiple dynamicrange states may be needed to achieve a preferred range for the RFIDsystem. For example, from about 1 mV to 1 volt is approximately a 60 dBbaseband dynamic range, but only about 30 dB RF dynamic range in squarelaw mode.

To solve this problem, multiple dynamic range states may be implemented,or deliberate compression as above, or both. Typical RF dynamic rangerequirements are from approximately +20 dBm down to a noise limited tagsensitivity level of approximately −60 dBm (bipolar) or −55 dBm (CMOS)without RF LNA. As discussed earlier, with an LNA it is possible toextend sensitivity to be as good as approximately −80 to −90 dBm.However, one skilled in the art will recognize that embodiments of theinvention cover any range and/or combination of dynamic range stateswithin an RFID system.

FIG. 7 illustrates a more advanced small signal model of the transistordetector including noise sources according to various embodiments of theinvention. This model may be used to analyze noise and the resultingsignal-to-noise ratio within the tag, leading to analytic sensitivitypredictions. Though the discussion below is often in terms of a bipolardetector transistor, this same noise model also applies to CMOStransistor implementation. In fact, CMOS transistors in “weak inversion”have a very similar exponential transfer function to bipolar, and canmake excellent square law transistor detectors also. “Weak inversion” or“subthreshold” operation of the MOS transistor is that region ofoperation where the gate to source voltage difference is less than thethreshold voltage of the device. In that regime the first order model ofdrain current holds that drain current is zero. This is not completelytrue, as the current is only very small. In that regime the currentfollows an exponential relationship of drain current to gate to sourcevoltage, and acts much like a bipolar transistor as a square lawdetector. It is the use of weak inversion mode that allows CMOS to reachas low as −55 dBm without LNA. However, the weak inversion mode CMOStransistor does have greatly reduced bandwidth. Thus, as a square lawdetector, under some operating conditions it may be necessary torestrict its use to lower data rates, and then use strong inversion forhigher data rates. In strong inversion mode, the noise limited CMOSsensitivity without LNA for typical process noise parameters is on theorder of approximately −50 dBm, which is still an excellent sensitivityfor RDID tags. Thus, the discussion below is not intended to be limitedto bipolar detectors, but to apply to MOS transistor detectors andsupporting circuitry also.

An RF input 700 with peak voltage V_(s) is AC-coupled via resistor 702and capacitor 704 resulting in a detected current ΔI_(b). A bias voltage710 is generated which includes noise voltages 712 and 718 generated bybiasing elements within the detector circuitry as well as noise frombaseband elements within the transistor detector itself. These are shownas resulting in a voltage bias and noise Vin_(n) 732.

If R_(bias2) 720 is much larger than R_(in) 708, then the detectedcurrent ΔI_(b) will almost exclusively flow within the base of thetransistor. As a result, the current through the base is totally oralmost totally sensed by the transistor and may be multiplied by thegain β_(t) of the transistor which allows for a much more efficientdetection of a weak RF signal when compared to the diode-based detector.

Current I_(b)+ΔI_(b) 726 is increased by the multiplication or gain of acurrent 728 which represents the gain of the transistor. Input noiseVin_(n) 732 is multiplied to output noise shown as I_(n1) 730, whichrelates to the transconductance of the gain circuit in the biascondition used. Additional output side noise current I_(n2) 734 is dueto inherent noise current on the output, such as flicker noise in thetransistor. Flicker noise is a particularly limiting factor in CMOSimplementation, and its effect is suppressed by the disclosed PNsequence methods according to various embodiments of the invention, asthis PN sequence methods allow higher frequency AC coupling that appliesa filtering effect to the low frequency flicker noise. An output voltage738 is produced across load resistor 736.

A derivation of the tag receiver sensitivity is now provided below usingthis model. Referring to the detector transistor, a change in collectorcurrent in relation to a change in input signal RF voltage is given by:

$\begin{matrix}{{\Delta\; I_{c}} = {\frac{I_{c}V_{s}^{2}}{4V_{T}^{2}} = \frac{I_{c}P_{in}Z_{in}}{2V_{T}^{2}}}} & {{Equation}\mspace{14mu} 23}\end{matrix}$In this equation V_(s) is the peak of the RF input sine wave signal. Ifthe input capacitance is not resonated, then the input impedance may berepresented by the following equation:

$\begin{matrix}{Z_{in} = {Z_{{Cin}\;}\;{Parallel}\mspace{11mu} R_{in}}} & {{Equation}\mspace{14mu} 24}\end{matrix}$However, if the input capacitance is resonated, the input impedance isincreased and may be represented by the following equation:

$\begin{matrix}{Z_{in} = {R_{par}\mspace{11mu}{Parallel}\mspace{11mu} R_{bias}}} & {{Equation}\mspace{14mu} 25}\end{matrix}$

In the above equation, R_(par) is the effective parallel resistance duethe limited Q of the resonant circuit that resonates out effective inputcapacitance C_(par).

As previously discussed, the receiver may be very sensitive to noise ifoperating in certain modes. The total output side noise may berepresented by the following equation:

$\begin{matrix}{{TotNoise} = {{N_{F} + {GainedNoiseIn}} = {N_{F} + {g_{m}^{2}{Vin}_{n}^{2}R_{L}}}}} & {{Equation}\mspace{14mu} 26}\end{matrix}$where:

-   -   Vin_(n)=detector input side noise voltage over the bandwidth of        interest    -   N_(F)=detector output side noise power over the bandwidth of        interest    -   R_(L)=detector load impedance    -   g_(m)=transistor baseband transconductance (transfer function        from input small signal voltage to output small signal current)        Additional terms to be used are:    -   D_(R)=data rate    -   R_(n)=effective noise resistance on input side that would        generate “Vin_(n)” (often only slightly larger than R_(bias2)        720)    -   F_(f)=“Filter Factor” for a filter 3 dB bandwidth generally        moderately greater than bit rate D_(R) needed to pass the        signal, usually between 1.5 and 2.0 to allow fairly sharp edges        for simple slicer circuitry.    -   N_(bwf)=effective noise bandwidth factor increase greater than        filter 3 dB BW (F_(f)*D_(R)). This factor N_(bwf) is        approximately 1.22 for second order filtering, and 1.57 for        first order. Physically, this factor allows capturing the noise        passed by the filter outside its 3 dB bandwidth due to the fact        that the filter skirts are not perfect. Total Noise        BW=D_(R)*F_(f)*N_(bwf) and conservatively is approximately 3        times the data rate.

Using these definitions and equation, and assuming that electronicdevice noises have been filtered down to the thermal noise level (thatis the purpose of C_(bypass) 716) the input noise voltage may berepresented by:

$\begin{matrix}{{Vin}_{n} = {\sqrt{4{KTBR}_{n}} = {\sqrt{4{KTF}_{f}N_{bwf}D_{R}R_{n}} \approx \sqrt{12{KTD}_{R}R_{bias}}}}} & {{Equation}\mspace{14mu} 27}\end{matrix}$In the above equation, “R_(bias)” is the total “effective” noiseresistance combination of R_(bias1) 714, R_(bias2) 720, and any biascircuitry noise. If the low pass corner formed by C_(bypass) 716 and thetotal resistance is well below the signal passband, then R_(bias) iswell approximated by R_(bias2) 720. The last approximation on the rightof Equation 27 applies if total effective noise resistance on the inputside is dominated by R_(bias) and if the total noise bandwidth isapproximately three times the data rate.

The useful baseband detected power for any transistor operating as asquare law detector is:

$\begin{matrix}{{SignalPwr} = {\frac{1}{2}\Delta\; I^{2}\mspace{11mu} R_{L}}} & {{Equation}\mspace{14mu} 28}\end{matrix}$Therefore, the signal to noise ratio on the output side of thetransistor detector is:

$\begin{matrix}{{SNR} = \frac{\frac{1}{2}\Delta\; I_{c}^{2}R_{L}}{N_{F} + {g_{m}^{2}{Vin}_{n}^{2}R_{L}}}} & {{Equation}\mspace{14mu} 29}\end{matrix}$Solving for ΔIc_(req) as harvested collector signal current that isrequired to achieve a desired SNR:

$\begin{matrix}{{\Delta\;{Ic}_{req}} = {\sqrt{\frac{2\mspace{11mu}{SNR}_{req}}{R_{L}}\left( {N_{F} + {{Vin}_{n}^{2}g_{m}^{2}R_{L}}} \right)} = \sqrt{2\mspace{11mu}{{SNR}_{req}\left( {I_{NF} + {{Vin}_{n}^{2}g_{m}^{2}}} \right)}}}} & {{Equation}\mspace{14mu} 30}\end{matrix}$where I_(NF) is detector output side noise current.

If the output side transistor noise floor N_(F) is negligible, then:

$\begin{matrix}{{\Delta\;{Ic}_{req}} \approx \sqrt{2\mspace{11mu}{SNR}_{req}\mspace{11mu}{Vin}_{n}^{2}g_{m}^{2}}} & {{Equation}\mspace{14mu} 31}\end{matrix}$

For a particular transistor and biasing circuit, expressions forΔI_(req) and Vin_(n), may be inserted and solved for sensitivity. Thus,equating for harvested signal ΔI_(c) as a function of signal power(moved to RF voltage via impedance environment) from Equation 23 andΔIc_(req) from Equation 31 gives:

$\begin{matrix}{\frac{I_{c}P_{in}Z_{in}}{2V_{T}^{2}} = \sqrt{\frac{2\mspace{11mu}{SNR}_{req}}{R_{L}}\left( {N_{F} + {V_{in}^{2}g_{m}^{2}R_{L}}} \right)}} & {{Equation}\mspace{14mu} 32}\end{matrix}$

In Equation 32 I_(c) is standing bias current. Now solving forsensitivity as minimum input power P_(in) that gives required SNR:

$\begin{matrix}{P_{in} = {{Sensitivity} = {\frac{2V_{T}^{2}}{I_{c}Z_{in}}\sqrt{\frac{2\mspace{11mu}{SNR}_{req}}{R_{L}}\left( {N_{F} + {V_{in}^{2}g_{m}^{2}R_{L}}} \right)}}}} & {{Equation}\mspace{14mu} 33}\end{matrix}$

From Equation 33 it is found that bipolar transistor detectors withexcellent output side noise floors N_(F) (those not degraded by morethan a few dB above thermal noise) can achieve sensitivity on the orderof approximately −60 dBm. It is also noted that greater bias currentshould improve sensitivity as long as the transistor output side noiseis dominant over input side thermal or device noise. But, if gainedinput side noise is dominant, then sensitivity is not a function of biascurrent because g_(m) is proportional to bias current and the biascurrent functional dependence will cancel.

FIG. 8 shows an exemplary curve 804 of bipolar transistor sensitivity(in dBm) 802 as a function of bias current (in μA) 800 in thetransistor. It is obtained from Equation 33 using representativetransistor parameters, and represents a case where transistor noise isnot negligible but also is not severe. In this particular example,Z_(in) is equal to 200Ω, R_(b) is equal to 8 kbps, and R_(bias) is equalto 1.8 kΩ. This curve shows the relationship between sensitivity andcurrent such that an increase in the bias current results in an increasein sensitivity up to a point.

CMOS detectors are generally on the order of 5 to 10 dB inferior to thebipolar sensitivity due to moderately smaller signal harvest and thehigher flicker noise currents on their output side resulting in largernoise floor N_(F). The use of Manchester modulation and the PN sequencetraining, flagging, and synchronizing method result in precisionreference control and flicker noise rejection that can extract the bestpossible sensitivity from CMOS.

To allow for this excellent sensitivity using either bipolar or CMOStransistor detectors on the tag, a plurality of dynamic range states areprovided within the RFID system design disclosed herein.

F. Dynamic Range States within an RFID System

An RFID tag is provided with a plurality of dynamic range states inorder to cover a wide range from very close to the reader to the longerrange in accordance with the tag sensitivity disclosed above. Asdiscussed above, the SNR within the tag relates to the transmit power ofthe reader and the distance between the reader and the tag, both whichare important factors in the received power level at the tag.

FIG. 9A illustrates an exemplary chart of dynamic range states thatallow a tag to operate over a relatively longer received power rangeaccording to various embodiments of the invention. A first dynamic range910 is shown comprising a relatively higher received power and relatingto a first mode of operation for the tag. In this first mode, the tagmay function in a peak detection mode over all or part of the higherdynamic range state.

A second dynamic range 920 is shown comprising relatively lower receivedpower and relating to a second mode of more sensitive operation for thetag. In this second mode, the tag may function in a square law mode inorder to properly detect and process the received weak RF signals.

In certain embodiments, a protocol allows for multiple forwardmodulation modes that the more sensitive dynamic range state may be usedfor a superior modulation mode such as Manchester, while the upperdynamic range state also serves well for the standard passive forwardmodulation form of Pulse Interval Encoding (PIE) in addition toManchester. One skilled in the art will recognize that the variousdynamic ranges may vary in characteristic and power according to theenvironments in which they operate.

Referring now to FIG. 9B, detector final output voltage 960 is shown asa function of two similar dynamic range states 964 and 968. An overlaprange 966 is shown between the low dynamic range 968 and the highdynamic range 964 that reduces the odds that the signal may be lost dueto fade events. Accordingly, when P_(in) 962 lies within the overlaprange 966, the tag may operate within either of the low or high dynamicrange states.

G. Signal Training and Synchronization

As previously discussed, embodiments of the RFID system provide for tagsto be operable in a plurality of dynamic state ranges. A transition timemay be provided within the system to allow the tags to completelytransition from one dynamic range state to another. In variousembodiments of the invention, a training period is provided withinactivation and normal mode command structures that allow this transitiontime. In many embodiments, this training period is important due to thewide ranges of desired signals present within the environment in whichthe RFID system operates. For example, if a signal is too strong or hasbecome too strong since the last dynamic range adjustment, then thedetector of the tag may saturate. In that case, with proper trainingtime provided, the BAT has enough time to detect the aforementionedcondition, and move to a less sensitive dynamic range state. On theother hand, if the desired signal is weak, the BAT may move to the moresensitive dynamic range state.

In yet another embodiment of the present invention, by means of an AGC,the dynamic range can be adjusted or extended within each dynamic rangestate. The closed loop control of the active load that was previouslydiscussed is one such form of AGC. Additionally, deliberate compressioncan extend dynamic range performance, and has also been presented.

In addition, methods may be provided that allow for AC couplingtraining, or equivalently, for quasi-AC coupling or quasi-DC coupling(two forms of adaptable slicer reference acquisition, as discussedbelow). Use of well controlled comparator reference levels enable thisimprovement is sensitivity according to certain embodiments of theinvention.

H. Non-PN Sequence Based Methods and Basic Forms of Slicer ReferenceAcquisition

In many instances, it may be desirable for the tag to be able to process(slice) detected forward link signals that may have low amplitudes, suchas approximately one to a few millivolts, at a comparator (slicer)input. For a comparator to properly “square up” such low amplitudesignals, the signal swing should be nearly perfectly centered around areference voltage on the other comparator input. This process ofcentering is referred to as “slicer reference acquisition.” Amongtraining processes that may be employed in the tag receiver, referenceacquisition comes after dynamic range state acquisition and beforesymbol timing acquisition (bit synchronization).

Since it is common practice to AC couple signals with large DC contentand small AC content in order to perform this centering via strippingoff the DC content, it is also common to use the term “AC coupling” as asynonym for reference acquisition. However, speaking accurately, ACcoupling for this purpose is really only a type of referenceacquisition. Other types do exist, and there are several that arepertinent to RFID tag receivers. For example, if a tracking systemderives a suitable reference and then holds it stationary over acommunications interval of interest, this may be referred to as “DCleveling” or “quasi-DC” coupling. It is not pure DC coupling since thereference so acquired is only temporarily fixed.

Slicer reference acquisition or training of an AC-coupling device may beassisted at the protocol level by a symbol-based, CW-based, or PN(Pseudo-random Noise) sequence based training technique. FIG. 10Aillustrates possible Manchester symbol-based training by designating atime period in the preamble of a repeating set of Manchester bits thatallows a BAT receiver sufficient time to select an appropriate dynamicrange state according to various embodiments of the present inventionand then conduct reference acquisition. Manchester is suitable for smallsignals and high sensitivity as it may be AC coupled to strip off DC andallow subsequent processing to apply to the AC signal of interestbecause it is “DC balanced” whereby each symbol has identical DCcontent. This leads to a random Manchester bitstream having a powerspectral density that approaches zero at zero frequency. Thus, if ACcoupled or high pass filtered with an AC coupling corner (high passfilter 3 dB point) is used that is well below the main spectral content,it will be passed with little distortion.

In FIG. 10A, a time of 11 symbols 1002 is first communicated to thereceiver to allow proper dynamic range selection and any AGC actions. Anadditional 21 symbols 1004 are next communicated for training ofAC-coupling or other method of slicer reference acquisition. Each symbolwill have a particular length 1000 (here 125 μs for a slow hibernatemode communications), and the sum of the training symbol times used willdefine the period of time for training. One skilled in the art willrecognize other time periods and symbol counts may be implemented in thepreamble of a BAT to generate a sufficient training period.

FIG. 10B illustrates an exemplary method of “Advanced PIE” pulseinterval encoding forward modulation preamble training according tovarious embodiments of the invention. “Advanced” PIE in differs from“Simple” PIE in that it has hibernation and some additional commands. Byproviding training of the tag receiver prior to transmission of theActivation command, the tag may be designed to capture a demodulationreference level for data slicing via a zero power RC low pass filter orother low power linear method, in preference to power consumingnon-linear active modes of reference acquisition. Such non-linearmethods (such as quasi-DC to be presented shortly) may be acceptable orpreferable once the tag is activated, since it is in normal or awakenedmode only briefly and thus with little impact on tag battery life. Butin hibernate mode, where the tag may be continuously listening, it maybe preferred to reduce tag power consumption.

The preamble shown in FIG. 10B is suitable for AC coupling or referencetraining of hibernating PIE tags according to various embodiments of theinvention. However, one skilled in the art will recognize that thispreamble may be less suitable if the hibernation PIE symbol time wereever reused as a normal mode data rate. In that event, the statisticalprobability of confusing this preamble with normal mode commands anddata would significantly increase, resulting in power expenditure in thetag to check if normal communications were in fact Activation commandsto which the tag should react. This problem will be addressed in thenext section disclosing PN sequence based training, activation validity,and frame synchronization.

FIG. 10C provides an illustration of “True AC Coupling” according tovarious embodiments of the invention. The input voltage V_(in) 1050contains a time varying AC signal V_(sig) and a DC component V_(DC1).FIG. 10D shows input signal V_(in) “swinging around” V_(DC1). If at acertain time V_(sig) “comes on,” then after a settling time of a numberof time constants of the coupling capacitor 1052 and coupling resistor1054, the DC voltage between V_(ref) 1056 and the DC content of V_(in)will be charged across coupling capacitor 1052. At that time, the ACcomponent of V_(in) will be seen at the positive comparator input as“swinging around” comparator reference level V_(DC2). This is theprocess referred to as “AC coupling training”, and once it is complete,the comparator will accurately slice or “square up” the input signal andpass it on to digital circuitry for further processing. Since comparator1058 cannot begin properly squaring up the signal until this trainingprocess has occurred, time be provided to allow this training process tocomplete within an RFID system.

FIG. 10E illustrates “Quasi-AC Coupling” whereby the DC content of theinput signal is “filtered out” by the RC filter and used as a comparatorreference according to various embodiments of the invention. The RCcircuit is in a low pass form as compared to the high pass form of trueAC coupling, but the training time requirements are typically the same.It is a case of resistively charging a capacitor to either move the DCpresented to the comparator input to match a fixed reference (true ACcoupling), or moving comparator reference to match the DC content of thesignal (quasi-AC coupling).

FIGS. 10F and 10G show the more sophisticated case of “Quasi-DCCoupling” according to various embodiments of the invention. This issimilar to quasi-AC coupling in that the comparator reference is beingmoved to center up for slicing. However, the comparator reference ismoved by an intelligent digital or non-linear circuit illustrated hereas variable voltage reference 1082 that acts upon or processes the inputsignal to obtain the desired reference level. One form of suchprocessing would be to use peak detectors, capture the positive andnegative peaks of V_(in) 1080, and obtain the average of those as areference level for slicing. One skilled in the art will recognize thatvarious methods may be employed to obtain the desired reference level;all of which are intended to fall within the scope of embodiments of theinvention.

There are several advantages of quasi-DC reference acquisition, such asfaster acquisition that can allow a lower training time. Anotheradvantage is that the reference voltage may be temporarily “frozen” asillustrated in FIG. 10G. In particular, V_(ref) 1082 allows quasi-DC toperform accurate slicing of modulation forms such as PIE that do nothave a 50% DC content, or even a fixed DC content. The waveform ofV_(ref) 1082 is shown as quickly adapting to the input signal andacquiring a reference quite close to the true midpoint of the signalswing of V_(in), with only a small “tracking error.”

This ability to freeze the slicer voltage reference for a desiredinterval of communication is referred as “quasi-DC” and relates to ageneral method of reference acquisition according to certain embodimentsof the invention. However, quasi-DC has the disadvantages of morecomplexity, cost, and power consumption as compared to AC or quasi-ACcoupling. In a hibernating RFID tag, where low hibernation powerconsumption is important to getting a battery life of months or yearsout of a tiny battery, the power consumption of quasi-DC referenceacquisition may be unacceptable.

Hence, various embodiments of this invention may use sufficiently longsymbol training in hibernating PIE mode to allow for low power AC orquasi-AC reference acquisition, only reverting to quasi-DC for the briefperiod of time the tag is awake in normal mode. When the PIE tag is innormal mode, specific symbols (such as RTcal and TRcal within theISO\IEC 18000-6C standard) may be used to communicate forward andreverse data rate information from reader to tag. These symbols are muchlonger than the standard zero and one symbols, have DC content thatvaries drastically from 50%, and yet still must have their pulse widthsaccurately measured by the tag to determine these data rates. However,the extreme duty cycle of the these long symbols oftentimes tends topull off AC coupling or quasi-AC coupling from the desired nearmid-point values that would allow accurate measurement, particularly atthe lower sensitivity levels desired of battery tags.

The problem of maximizing sensitivity of battery tags using thenon-ideal PIE modulation mode may be partially addressed by acquiringthe reference with only the zero and one symbols in the hibernate mode,and then for the case where the tag is the first accessed after wakingup, temporarily “saving” the reference using quasi-DC modulation to useduring the time that the long symbols (e.g., RTcal and TRcal) need to bemeasured in normal mode. If a tag is not immediately accessed in normalmode, then during the time that the reader is accessing other tags, thetag may “refresh” its slicer voltage reference periodically by theavailable normal reader signaling and be ready to accurately sliceforward reader communications when the tag's turn to communicate doesarrive.

FIG. 11 illustrates a CW-based training technique according to variousembodiments of the invention. A system may implement the CW-basedtraining 1100 because it has only one spectral component (at the carrierfrequency), that allows the CW-based training to create lessinterference by not transmitting frequency components in other channelsthat could interfere with the relatively weak tag backscatter signalsthat other readers are receiving. In this example, a CW is maintained at50% peak power (i.e., DC average of the signal) or approximately 50%peak power for a period of time 1102 that allows sufficient trainingtime for the BAT.

FIG. 12 is a plot 1200 that illustrates the advantage of CW trainingover symbol training based on the spectral occupancy of the differenttraining methods. The plot shows frequency power (in dB) 1204 as afunction of the frequency of the signal 1206. In this example, arepeating Manchester symbol preamble (all ones) is shown within thefrequency domain at which the carrier 1208 is shown in the middle andother frequency components on both sides of the carrier. Comparatively,CW training causes a single impulse at the sine wave carrier frequencyin the frequency domain. Accordingly, the CW carrier causes a lowerstatistical incidence of interference during the training operations andmay be preferred in many environments in which frequency spray mayadversely interfere with the relatively weak backscatter signals thatother readers are receiving. One way CW training may be advantageouslyused is to allow a CW interval to precede the PN training sequences thatare disclosed herein.

I. PN Sequence Based Training and Synchronization

The above training methods using series of all ones or zeroes, or simplypure carrier, have the advantage of simplicity, but may not provide thebest possible total performance relative to all the desired functions ofa preamble. For example, in order to distinctly differentiate Activationcommands from other commands or data at the same data rate, prior artmethods used longer symbols as flags that a legitimate activation wasbeing attempted. Such long symbols, being distinct from the normal zeroand one symbols, could also serve as frame synchronization flags.

Frame synchronization is generally the next synchronization requiredabove symbol or bit synchronization and is used to process thebitstream, such as allowing the bitstream to be parsed into words orcommands. An example of a symbol that may serve as a frame marker or“flag” is a symbol noticeably longer than the zero and one symbols, suchthat it is clearly distinguishable from the zero and one symbols. AnRFID industry term for such a long symbol frame flag is an “interrupt”.However, interrupts have the disadvantage of requiring AC couplingmethods that have lower high pass corners, which are more expensive ofintegrated circuit die area and take longer to train. The lower ACcoupling corner that passes these interrupts also allows more lowfrequency flicker noise in the demodulator, and may noticeably harm tagsensitivity, particularly in CMOS implementations. In many RFID systems,it may be preferred to perform all authentication and synchronizationfunctions using only the standard symbol alphabet, which for RFID isnormally the zero and one symbols, because higher level signaling (suchas 2 bits per symbol, requiring 4 symbol states) is not common practicefor the forward link of simple and low cost RFID tags.

Referring now to FIG. 13, an implementation of a hibernation modeActivation command preamble using PN sequences is shown according tovarious embodiments of the invention. This preamble consists of a set ofPN sequences that achieves the combination of dynamic range selection,AC coupling or reference acquisition training, bit synchronization, andframe marking. In various embodiments, the frame marker is an importantcommunication function and in the advantageous form shown here may alsoprovide the additional function of a valid “activation flag” oractivation authentication to indicate to a high reliability that alegitimate Activation command is about to follow. This not only providestraining, symbol synchronization, and frame synchronization at the tag,but allows the tag to begin powering up circuitry to process thefollowing Activation command and prepare for normal mode operation ifthe activation is successful. This preamble may also be enhanced with apreceding period of reader CW transmission that allows additionaltraining time for dynamic range selection while causing lowerinterference through its very narrow spectrum.

In many situations, it may be advantageous to provide the tag the optionto begin its power-up sequence as early as possible, so that time may besaved for the power up of key tag subsystems. Examples of such keysubsystems that have noticeable time lag associated with their power updelays include the start up times of crystal oscillators and phaselocked loop synthesizers, start up routines of microcontrollers, andsettling times of voltage regulator based power supplies. If the tagengages in too high a density of these start ups, only to later shutdown because it turns out that there was not a legitimate activationunderway, then its battery life may be significantly shortened. The highreliability of the frame marker flag prevents the tag from depleting itsbattery with at least some unnecessary processing.

The particular set of PN sequences in the activation preamble of FIG. 13start with a sequence of 21 bits 1300 selected out of a 31 bitm-sequence (a type well known in the art). The first 7 bits 1304 areintended for dynamic range state selection and AGC action (ifnecessary), and the following 14 bits 1308 for AC coupling training orother method of slicer reference acquisition, and symbolsynchronization. An advantage of a 31 bit m-sequence is that thispreamble may in the future be conveniently extended to the full 31 bits.One reason this may be desired is for selection between more than twodynamic range states, such as might be desired for the case of an RF LNAused in a Class 3 Plus tag, resulting in 3 or 4 dynamic range states.Following the 14 reference and symbol synchronization bits, FrameMarker/Activation Flag 1310 consisting of all 15 bits of a 15 bitm-sequence 1302 is transmitted by the reader. The addition of bitstuffing in the reader and destuffing in the tag that this 15 bitm-sequence frame marker flag is reliable as a frame synchronizer andindicator of legitimate activation, even in the presence of bit errorsin the channel.

The preamble of FIG. 13 concludes with 2 “one” bits 1312 to provide afinal timing trim and preparation to receive opportunity to the tagprior to the subsequent full Activation command. This period of time maybe crucial to having adequate power-up sequencing time for tags that didnot take advantage of the opportunity to begin power-up sequencing uponpartial correlation of the frame synchronization/activation flag thatpreceded the final timing adjustment. One skilled in the art willrecognize that various hibernation mode Activation command preambles maybe used with embodiments of the invention.

Referring now to FIG. 14, an exemplary normal mode command preamble isshown according to various embodiments of the invention. This preambleis similar to the preferred activation mode preamble of FIG. 13. Itstarts with same 21 bit 1400 truncated 31 bit m-sequence of FIG. 13 forthe various training steps of dynamic range state selection (if used),Automatic Gain Control (if used) AC coupling/reference acquisitiontraining, and bit synchronization. Specifically the first 7 bits 1404are advantageously used for dynamic range and/or AGC adjustment, and thenext 14 bits 1406 for AC training or slicer voltage referenceacquisition. It then has a 15 bit frame marker 1408 constructed ofm-sequence 1402 that may advantageously be the bit by bit logicalinverse of the 15 bit frame marker/activation flag 1310 of FIG. 13.Using the bit inverse of the same frame marker allows reuse of framemarker detection circuitry on the tag. The preferred normal modepreamble of FIG. 14 differs from the preferred hibernate mode preambleof FIG. 13 in that it does not have a two bit timing adjust on the endof the preamble. Just as in the case of the hibernate mode preamble, itsoperation may be enhanced by a period of CW that precedes the firstmodulated PN bits. Such preceding training is most advantageouslyconducted at 50% of the peak power to be used during modulation, whichmeans this CW then accurately represents the average power to be usedduring modulation. Since it is equal, it allows dynamic range selectionand reference acquisition training. This then allows the PN bits thatprecede the frame marker to all be used for symbol synchronization,which can allow for more accurate bit synchronization methods.

Simplification of the PN correlator circuitry on the tag may also bepossible. In certain embodiments, a series of shorter PN sequences(subsequences) are used to make up a longer total sequence. Thecorrelator may be simplified to use the shorter PN sequence over andover, for example, incrementing a counter value upon each successfulcorrelation. Full correlation is designated when every subsequence hasbeen successfully correlated.

In various embodiments, the reader may adjust the subsequence length andnumber of successfully correlated repeats needed to declare successfulcorrelation. This flexibility allows trade-off of the statistics ofreliable correlation required vs. hostility in the channel. For example,for very hostile channels where it may be difficult to pass thecorrelation criteria, the reader may establish relaxed criteria. The tagmay also, based upon internal criteria, be allowed to adjust theseparameters independently of the reader, for example, when the tagdetects a situation where it is receiving what appear to be multipleattempts to send an Activation Flag that are missing correlation due toa high Bit Error Rate on the channel.

This adjustment of parameters might also be useful if the hibernate modesensitivity is degraded as compared to the higher power consumptionnormal mode sensitivity. For example, a Class 4 tag will have an activereceiver that is normally more sensitive and more selective than itshibernate mode square law receiver. A channel that is difficult for thetag's square law hibernate receiver may be perfectly usable for itsactive mode receiver, if the reader can get the tag to go active andturn that better receiver on, if the tag detects activity that itsuspects is RFID and then autonomously turns on its active receiver tocheck. These considerations of adaptable threshold of correlation alsoapply to the case of a single longer PN sequence. It may also beadvantageous to store these variable operational settings in theSettings File to be discussed later with regards to the Tag CapabilitiesReporting and Setting system.

Part of the reason to use PN sequences as frame synchronization flags isthat they offer good cross-correlation properties. In the case of asimple digital correlator, which may be implemented on a tag, thebit-by-bit sum of two versions of the PN sequence, where one is shiftedat least one bit differently than the other, has a low sum value. A fullsum (15 for a 15 bit sequence) only occurs when the shifts are equal,which reliably indicates synchronization. The correlation of sequenceformed of part of a PN sequence plus some random (noise) bits againstthe full sequence is also generally low.

For example, FIG. 15 presents a realistic cross-correlation 1500 of asequence of the 21 NRZ training bits followed by the 15 bit m-Sequenceflag presented in FIG. 14 in the case where the receiver correlator waspre-loaded with random data (for example when the tag is powered up).Once the transmission begins, the demodulator is temporarily over-loadedduring initial training (the receiver may not be in the correct dynamicrange state and the AC coupling charge state does not provide thecorrect slicing reference), as the initial random data in thecorrelator's buffer is shifted out, the wrongly sliced data is shiftedin until the point the demodulator begins to correctly demodulate (atthe 37th Manchester half-bit). It can be seen that the value of thecorrelator's output does not get close to its peak until the full 15-bitPN flag is completely in the buffer. This is the advantage of using a PNflag, when it is correlated to any other signal (including itself butshifted, even slightly shifted), the value of the correlator's outputremains low, until the whole sequence is in the correlator's buffer (atthe 72nd Manchester half bit), at which point the correlator's outputpeaks, reliably signaling correct frame synchronization.

In the above discussed exemplary PN sequence preambles, 15 bits of PNsequence for activation validity flags and frame synchronization wasdisclosed. The system design concern is primarily to prevent falserecognition of PN sequence frame markers and activation validity flags.There are various problems that result from such false recognitions asfollows:

-   -   In normal mode operation, if normal data traffic is mistaken by        the tag for a PN frame marker, then the tag may subsequently        mistake the following bits as a command that it could mistakenly        execute.

In hibernate mode, if normal data traffic is mistaken by the tag for aPN frame marker/activation validity flag, then the tag may subsequentlybegin its wake up process, thus wasting battery life.

In various embodiments of the invention, bit stuffing may be implementedin the reader data streams so that PN sequence flags are notaccidentally transmitted. In certain embodiments, the flag length isN_(flag) bits long. On the transmit (reader) side, when the readerdetects it will transmit a forbidden sequence (i.e., 14 bits(N_(flag)−1) match the sequence), the reader inserts one additional bit,effectively forcing a different sequence. On the receive side, theinverse process occurs. When the tag detects it is on the brink ofreceiving the forbidden sequence (the first N_(flag)−1 bits match), itexamines the last (15^(th)) bit to determine if it is a stuffed bit (itis if it is different from the last bit of the PN sequence). If it is astuffed bit, it is removed from the bit stream. If it is not differentfrom the last PN bit, it is a valid PN flag and is not unstuffed.

FIG. 16 shows a method of “unstuffing” in the tag according to variousembodiments of the invention. Demodulator 1602 is providing demodulatedbits to be evaluated for PN sequence validity by correlator and stuffedsequence detector 1608. In certain embodiments, these two functions arecombined into a single module and in other embodiments the functions areprovided in discrete modules. Data from demodulator 1602 issimultaneously shifted into correlator 1608 and shift register 1614.When a stuffed bit is detected, indicating there is not a correct PNsequence but a stuffed data stream that almost correlates, thencorrelator 1608 interrupts the clock of shift register 1614 for a singlecycle. This removes the stuffed bit from the serial bit stream,resulting in destuffed bit stream output 1616.

J. PN Flag False Command Response on Real Data

In the absence of bit errors, stuffing has prevented False CommandResponse due to a tag misinterpreting reader data other than commands onthe channel. A False Command Response can occur on real transmissions ifthe decoder misses the command and a bit error occurs in the readerstuffed bit stream. That situation can occur in a usable but still errorprone channel (about 1E-2 to 1E-4 BER), but even then there isstatistical protection from the length of the flag code. Most longercommands, such as those within the ISO/IEC 18000-6C RFID standard, alsohave CRC protection, which allows successfully limiting the FalseCommand Response that can occur when longer commands with parametersand/or data could emulate the flag plus a shorter command that is notCRC protected. In ISO/IEC 18000-6C, Manchester commands that are notcurrently CRC or specifically random number “handle” protected areQuery, QueryRep, QueryAdjust, NAK, and BroadcastID. The worst case isthe very common QueryRep command, which is only 4 bits long and thusstatistically likely for random data to emulate. The command most likelyto suffer a bit error that could falsely emulate a PN flag and theQueryRep command is the Select command, due to its possible long length.

A False Command Response with actual reader transmissions can only occurif these conditions are fulfilled:

-   -   The tag misses the command, since when the tag gets the command        it knows to lock out the specific length of known command        following bits from any misinterpretation as a flag    -   A bit error occurs in a critical place to turn a sequence that        is one bit different from the flag into the flag (the situation        of a string that is more than one bit different being turned        into the sequence is possible, but statistically negligible for        workable error rates)    -   The bits after the false flag match up to those expected for the        command

The total False Command Response probability is approximately theprobability of missing the flag times the probability that a bit errorhappens in one of the strings that can be turned into the flag sequencetimes the probability of a match to the 4 or more bits necessary toemulate a command.

Next the following definitions are made:

N_(flag)=the flag/frame synch length in NRZ bits

C_(CL)=the command code length (4 for Select)

E_(d)=length of essential data such as length (7 bits in the Selectcommand)

C_(L)=total command length (code+all parameters and data)

P_(e)=probability of bit error

P_(pcfr)(Command)=Probability of Particular Command False Response

The probability of missing the frame synch flag plus the command is theprobability of at least one error in the flag, command code bits, andthe parameters necessary to know the length of the command. If a normalmode frame synch flag size of N_(flag) NRZ bits is used, then:

$\begin{matrix}{{P\left( {{Missed}\mspace{11mu}{\_ Command}} \right)} = {1 - \left( {1 - P_{e}} \right)^{N_{flag} + C_{CL} + E_{d}}}} & {{Equation}\mspace{14mu} 34}\end{matrix}$

In a channel that may be modeled as “All White Gaussian Noise” (AWGN)limited:

$\begin{matrix}{P_{e} = {0.5 \cdot {{erfc}\left( \sqrt{SNR} \right)}}} & {{Equation}\mspace{14mu} 35}\end{matrix}$

In this equation, “erfc( )” is the complementary error function wellknown in the communications systems design art, and SNR is the signal tonoise ratio in linear units.

Next, the mean number of strings in the command is calculated that areone bit off from the flag sequence. For flags that have N_(flag) bits,this consists of those strings that had N_(flag)−1 first bits thatmatched and were stuffed with the opposite final bit, and also thosestrings that out of N_(flag) bits just happen to have any N_(flag)−1bits that match. The strings of N_(flag) length that have N_(flag)−1matches are the dominant “almost matches” that a single bit error canmake into a match. The number of such events is attained via a standardBernoulli Trial calculation where the probability of the number ofoccurrences “k” of an event A with probability “p” of A in N trials is:

$\begin{matrix}{{P\left( {A\mspace{14mu}{occurs}\mspace{14mu}{exactly}\mspace{14mu} k\mspace{14mu}{times}\mspace{14mu}{out}\mspace{14mu}{of}\mspace{14mu} N} \right)} = {\frac{N!}{{k!}{\left( {N - k} \right)!}}{p^{k}\left( {1 - p} \right)}^{N - k}}} & {{Equation}\mspace{14mu} 36}\end{matrix}$

If the event is a bit match (with p=0.5) to occur N_(flag)−1 times outof N_(flag) bits, then the above equation gives:

$\begin{matrix}{{P\left( {{Nflag} - {1\mspace{14mu}{matches}\mspace{14mu}{before}\mspace{14mu}{stuffing}}} \right)} = {{\frac{N_{flag}!}{{\left( {N_{flag} - 1} \right)!}{(1)!}}\left( \frac{1}{2} \right)^{({N_{flag} - 1 + 1})}} = \frac{N_{flag}}{2^{N_{flag}}}}} & {{Equation}\mspace{14mu} 37}\end{matrix}$

In addition to the matches that occur in N_(flag)−1 bits that are notstuffed, there is also the single additional exact match of all bitsthat was stuffed. Thus the total odds of a string in the stuffed bitstring of N_(flag) bits that have N_(flag)−1 bits matching the flag are:

$\begin{matrix}{{P\left( {{Nflag} - {1\mspace{14mu}{matches}\mspace{14mu}{after}\mspace{14mu}{stuffing}\mspace{14mu}{in}\mspace{14mu}{Nflag}\mspace{14mu}{bits}}} \right)} = \frac{N_{flag} + 1}{2^{N_{flag}}}} & {{Equation}\mspace{14mu} 38}\end{matrix}$

Thus, the total mean number of strings in a long string of C_(L)(command length including parameters and data) bits that could bechanged to the flag by a single bit error is:

$\begin{matrix}{S_{1} = {{\frac{C_{L}}{N_{flag}}\frac{N_{flag} + 1}{2^{N_{flag}}}} \approx \frac{C_{L}}{2^{N_{flag}}}}} & {{Equation}\mspace{14mu} 39}\end{matrix}$

For each of the S1 strings that are one bit off, a bit error in thesingle different bit could change the string to match the flag. Thesingle bit error probability is P_(e). The total probability of aparticular False Command Response (without CRC protection and for aparticular command) is the probability of getting the command wrongtimes S₁ (off by one strings) times the probability of for each “oneoff” string of getting all the matching bits right but also getting abit error in the single different bit, and then times the probability ofgetting the particular (non-CRC protected) command code, and istherefore given by:

$\begin{matrix}{{P_{pcfr}\left( C_{LE} \right)} = {{\left( {1 - \left( {1 - P_{e}} \right)^{N_{flag} + C_{CL} + E_{d}}} \right) \cdot \frac{C_{L}}{N_{flag}}}{\frac{N_{flag} + 1}{2^{N_{flag}}} \cdot \left( {1 - P_{e}} \right)^{N_{flag} - 1} \cdot P_{e}}2^{- C_{LE}}}} & {{Equation}\mspace{14mu} 40}\end{matrix}$

This approximation is valid so long as the error rate is low enough thatthe rate of “one off” strings being turned into complete matches is muchgreater than the rate of “two off” strings being turned into matches.The error rate could also be lower if the flag and command werecorrectly received but there was an error in the length received. Inthat case this equation gives an upper bound on error instead of actualerror. The variable C_(L) is the total command length, and C_(LE) is theessential bits needed in random data following a false flag tosuccessfully counterfeit a command. The worst case C_(LE) is 4, for theQueryRep command of ISO/IEC 18000-6C.

This probability may now be examined using Equation 40 for the worstcase in practice of a Select command emulating a QueryRep, which hasparameters as follows.

-   -   Select Command Code: 1010, 4 bits    -   QueryRep Command Code: 00XX    -   Select Command Maximum Size: Theoretically unlimited due to        unlimited EBV pointer, but a practical limit of 308 bits when        the address uses 2 bytes, and the length is set to 255

FIG. 17 depicts the resulting False Command Response vs. Bit Error Rate(BER) for 15 bit NRZ flag 1706 and 11 bit NRZ flag 1708. From thisfigure, it is noted that flags of length 15 will keep False CommandResponse below 1E-6 almost to BER of 1E-2. An 11 bit flag only holds1E-6 False Command Response up to about 2E-3.

K. PN Flag False Command Response on Noise

Embodiments of the invention may also be applied to a Manchester case(where the PN flagging is currently drafted into ISO/IEC 18000-6C) of ahighly sensitive receiver under weak signal conditions that is in normalmode and can “toggle on noise” in the absence of any reader signal. Thiscould occur when all nearby readers are temporarily silent, such as in atime coordinated system where readers are systematically controlled toprevent reader-on-reader interference. Let:

-   -   N_(flag)=NRZ length of the flag    -   C_(LE)═Number of essential command bits that must match for an        accidental response (4 for QueryRep)    -   M_(FFs)=Mean False Flags per Second    -   D_(R)=Data rate    -   P_(pcs)( )=Probability of a false Particular Command per Second        on noise toggling Then,

$\begin{matrix}{M_{FFs} = {{2^{{- 2}N_{flag}}{WordRate}} = {2^{{- 2}N_{flag}}\frac{D_{R}}{N_{flag}}}}} & {{Equation}\mspace{14mu} 41}\end{matrix}$

$\begin{matrix}{{P_{pcs}\left( {{Particular}\mspace{14mu}{{Command}'}s\mspace{14mu} C_{LE}} \right)} = {{M_{FFs}2^{{- 2}C_{LE}}} = {\frac{D_{R}}{N_{flag}}2^{{- 2}{({N_{flag} + C_{LE}})}}}}} & {{Equation}\mspace{14mu} 42}\end{matrix}$

Now the worst case of the unprotected QueryRep command with C_(LE)=4,data rate=8 kbps, and N_(flag)=15, and we findP_(pcs)(QueryRep)=1.94E-9. Thus, very high protection of False CommandResponse is provided in the noise toggling case. This is a result of theeffective doubling of the number of flag and command bits by theManchester coding.

L. Battery Life Considerations in PN Flag Length Selection.

In certain embodiments of the invention, a flag length of 15 bits (NRZ)may also meet battery life requirements. In the case of very low currenthibernate mode tags, improving and/or maximizing the battery life is animportant criterion that drives the PN sequence length up to 15 bits.

Battery life degradation due to false flags is very difficult to avoid(if a partial wake-up on received activation flag is assumed), but theeffects can be significantly reduced.

Negligible density of false wake ups may be defined as those causing 1%or less reduction in battery life, though the below mathematics may beapplied to other accepted battery life reductions. The effect is evenless in the case of higher hibernate powers used to attain betterreceiver sensitivity. In certain instances, tags can wake up orpartially wake up after a flag match, such as those tags that need“warning” in order to get a head start on powering up tag systems. Thosethat require a full activation mask match for wake up suffer a lowerrate of false wake up, and those that check the Activation CRC-16 beforeany wake-up suffer a still lower rate of false wake-up.

The cases are examined of real data and of a tag receiver that is“toggling on noise”, or operating at the noise threshold without“squelch” for maximum sensitivity. In these cases, a problematicscenario is when the tag is receiving a low strength but deterministicsignal from a reader, and accidentally encounters a flag in data throughan error that defeats the stuffing. In this case, a flag length of 15bits is preferable to keep battery life penalty below 1%.

First the following variable definitions are made:

-   -   F_(w)=Fraction of battery life wasted by false activations    -   R_(Aon)=Rate of accidental wake-ups (on) per second. One of        these applies to accidental wake-ups from data transmitted at 8        kbps, and another from noise    -   I_(hib)=Current draw during hibernate    -   I_(on)=Current draw during temporary wake-up. In certain        instances, a tag may not wake up unless there is a full mask        match, and others unless there is a full mask match and a        CRC-16, and these do not suffer a noticeable false wake up.    -   T_(Aon)=Time Accidentally awoken tag is on before detecting        accident and going back to sleep    -   D_(R)=Forward Data Rate    -   B_(DF)=Bits to Detect Failure (accidental wake-up,        conservatively 30, but perhaps as many as 168 if not detected        until CRC-16 at end of activation is checked)    -   N_(flag)=PN sequence flag length in NRZ bits    -   BLAF=Battery Life Acceptability Factor

The fraction of battery life wasted will be selected as a designparameter. Then, we may write by inspection that:

$\begin{matrix}{F_{w} = {\frac{AvgWaste}{Draw} = \frac{I_{on}T_{Aon}R_{Aon}}{I_{hib}}}} & {{Equation}\mspace{14mu} 43}\end{matrix}$

Equation 43 may be rearranged as an inequality to depict minimum Nflagas a function of fraction wasted F_(w):

$\begin{matrix}{\frac{F_{w}I_{hib}}{I_{on}T_{Aon}R_{Aon}} = {{BALF} \geq 1}} & {{Equation}\mspace{14mu} 44}\end{matrix}$

Equation 44 is interpreted to mean that if the quantity Battery LifeAcceptability Factor (BALF) is greater than 1, then the parameter sethas resulted in battery life fraction Fw of less than the chosen amount.

It may be noted that:

$\begin{matrix}{T_{Aon} = \frac{B_{DF}}{D_{R}}} & {{Equation}\mspace{14mu} 45}\end{matrix}$

$\begin{matrix}{\frac{F_{w}I_{hib}D_{R}}{I_{on}B_{DF}R_{Aon}} \geq 1} & {{Equation}\mspace{14mu} 46}\end{matrix}$

From this point the analysis diverges into the two cases of wake-upsfrom reader data and wake-ups from a wide open tag receiver toggling onnoise.

1. Case 1: Accidental Wake-Up Due to Deliberately Transmitted Data.

In certain embodiments of this invention, partial or entire correlationof the Activation and Frame Synchronization flag portion of the preambleis used to allow an early start to tag power-up sequencing, instead ofwaiting until the full Activation command is correctly received anddecoded. This can reduce delay time before the tag is ready to beginnormal mode operations. For example, it can take milliseconds to tens ofmilliseconds to power up a crystal oscillator and/or phase locked loopsynthesizer for a Class 3 Plus or Class 4 tag, or to power up and lock aclock synthesizer for a microcontroller clock for a Class 3 tag.Accordingly, the PN sequence Activation Flag is designed so that thisoptional early start on power-up sequencing does not accidentally happenoften enough to noticeably degrade battery life.

An analysis of this case makes use of the earlier development of thestatistics under which the bit stuffed channel can by way of errors inthe channel still cause an accidental start to wake-up. The channel willbe stuffed such that in an error free channel the wake-up flag does notoccur. However, in the presence of errors, the stuffed strings of lengthN_(flag) (which are off from a full match by one bit), and the otherstrings of length N_(flag) that happen to be off from a full match byone bit, can suffer a bit error that creates the flag. If the tag beginsits wake up process at that time (to power up sub-systems) instead ofwaiting for the mask and the Activation CRC, then the tag can for someperiod of time waste power before it detects a false wake-up and goesback to sleep. In this case, R_(Aon) (false flags per second) is givenby the FlagRate X FalseFlagOdds, which is given by:

$\begin{matrix}{R_{Aon} = {\frac{D_{R}}{N_{flag}}\left( {1 - P_{e}} \right)^{N_{flag} - 1}P_{e}\frac{N_{flag} + 1}{2^{N_{flag}}}}} & {{Equation}\mspace{14mu} 47}\end{matrix}$

Now substituting into Equation 46:

$\begin{matrix}{\frac{F_{w}I_{hib}N_{flag}2^{N_{flag}}}{I_{on}{B_{DF}\left( {1 - P_{e}} \right)}^{N_{flag} - 1}{P_{e}\left( {N_{flag} + 1} \right)}} = {{BALF} \geq 1}} & {{Equation}\mspace{14mu} 48}\end{matrix}$

Equation 48 is a test of sufficient length for N_(flag) to meet thedesired battery fraction wasted F_(w). Now we may examine particularcases of moderate sensitivity receivers of −40 dBm with hibernatecurrent of 100 nA with a long interval of 168 bits to determine that afalse flag occurred (worst case of full Activation command), and highsensitivity receivers of −55 dBm with hibernate currents of 2 μA and aquicker B_(DF) of 30 bits to detect a false flag. Variable parametersare defined as follows:

-   -   I_(hib)2 μA and 100 nA    -   F_(w)=0.01 (1% reduction in battery life)    -   I_(on)=25 μA (from wake-up to detection of false wake up)    -   B_(DF)=30 bits (time in bits to detect false wake up) for 2 μA        case and 168 bits for the 100 nA case    -   BER=0.01 (right at fringe of receiver sensitivity)

Referring now to FIG. 18, BLAF curve 1808 shows the 2 μA case. The curvecrosses the acceptable line of BALF>1 at N_(flag) between 6 and 7 bits.BLAF curve 1810 for 100 nA does not cross into acceptability until rightat 15 bits, hence is a strong reason to design and standardize N_(flag)at 15 bits.

2. Case 2: Accidental Wake-Up when Toggling on Noise.

By inspection,

$\begin{matrix}{R_{Aon} = {{({FlagRate})\left( {{Random}\mspace{14mu}{Flag}\mspace{14mu}{Probability}} \right)} = {{\frac{D_{R}}{N_{flag}}2^{{- 2}N_{flag}}} = \frac{D_{R}}{N_{flag}2^{2N_{flag}}}}}} & {{Equation}\mspace{14mu} 49}\end{matrix}$

Note that because N_(flag) is in NRZ format, there are two trials perNRZ bit to get a random matching Manchester bit, hence the factor of 2on N_(flag).

Substituting this relation into the basic equation for F_(w) (Equation43):

$\begin{matrix}{F_{w} = {\frac{I_{on}T_{Aon}R_{Aon}}{I_{hib}} = {{\frac{I_{on}}{I_{hib}}\frac{B_{DF}}{D_{R}}\frac{D_{R}}{N_{flag}2^{2N_{flag}}}} = {\frac{I_{on}}{I_{hib}}\frac{B_{DF}}{N_{flag}2^{2N_{flag}}}}}}} & {{Equation}\mspace{14mu} 50}\end{matrix}$

This equation may be rearranged as an inequality to use as an indicatorof acceptable length for N_(flag) for negligible false wake-ups. Whenthe below inequality exceeds 1.0, then the flag is of acceptable length.

$\begin{matrix}{{N_{flag}2^{2N_{flag}}\frac{I_{hib}F_{w}}{I_{on}B_{DF}}} = {{BLAF} \geq 1}} & {{Equation}\mspace{14mu} 51}\end{matrix}$

Now we may examine a particular cases of a high sensitivity receivers of−55 dBm with hibernate currents of 2 μA. Parameters assumed are:

-   -   I_(hib)=2 μA (case 1) and 100 nA (case 2)    -   F_(w)=0.01 (1% reduction in battery life)    -   I_(on)=25 μA (from wake-up to detection of false wake up)    -   B_(DF)=30 bits (time in bits to detect false wake up)

This result is graphed in FIG. 18, BLAF curve 1806, which satisfies thepractical cases examined, needing only N_(flag) of 7 to satisfy the 1%battery life reduction criterion. Thus, the worst case found was for the100 nA ultra-low receiver current case with false flags on real data,leading to a worst case N_(flag) requirement of 15 (curve 1810).

M. Extensible Command Structure and Interference Control

Interference control and future expandability are oftentimes importantrequirements of a wireless system. For example, most dense wirelesssystems such as cellular telephony are so troubled by interference thatthey are referred to as “interference limited” systems. Attainingacceptable functionality in such systems often revolves aroundinterference controlling system design. Unfortunately, RFID systems havenot historically been designed for effective interference control. Theirself-interference is fundamentally only acceptable because of thelimited sensitivity at both ends of the link.

The sensitivities disclosed herein thus lead to higher propensity forinterference that may be dealt with through deliberate system design.For example, the passive state machine behavior and command responses ofstandardized Class 1 RFID systems in the EPCglobal™ Gen 2 Class 1standard version 1.2.0, which are carried over into ISO/IEC 18000-6C,were designed assuming a much less sensitive tag than Advanced PIE andManchester allow. This can lead to certain non-ideal behaviors forbattery systems, such as the sensitive tags responding to any “Select”command (the command that pre-selects tags by attribute for inclusion ina subsequent operation called a Query round where they may be singulatedfor individual access) they receive from an “enemy reader” by departingthe state their associated reader desires them to be in and going backto the starting or “Ready” state.

To reduce the incidence of such undesired interference, Advanced PIE andManchester tags may provide an interference rejecting feature(s) ofoptionally not responding to commands with embedded Session flag unlessthose sessions match the session to which they were activated. (Thisstate is referred to herein as “Interference Rejection On”) This featureis further strengthened by the use of tag-to-reader locking whereby thetag receives an 8 bit reader ID code that even more reliably identifiesa tag's associated reader.

The reader may at its option activate the tag without Session Locking,also called “Promiscuous Mode”. “Promiscuous Mode” is typically onlyused in environments where there are low odds of interference from“enemy” readers (which under some circumstances is any reader other thanthe activating reader). Additionally, dense wireless systems usually usetransmit power level control to limit transmitted power to no more thanthat required to effectively conduct desired communications. Powerleveling may be advantageously added to RFID systems according tovarious embodiments of the invention by the preferred use of new commandset improvements and other features that enable this capability andwhich are described herein.

RFID and wireless systems oftentimes grow over time and may need toadapt to changes in technologies and standards. In the RFID case, aparticular growth opportunity is for higher performance battery assistedtags. In accordance with various embodiments of the invention, commandand feature sets are provided that facilitate this growth within an RFIDsystem and allow scalability across different RFID standards,technologies and devices.

In certain embodiments of the invention, a command structure is providedthat extends upon prior art passive RFID methods (such as those depictedwithin EPCglobal™ Gen 2 version 1.2.0 and ISO/IEC 18000-6C standards) toallow for much improved operation using battery assisted tags. Examplesof such improvement include:

-   -   “Tag-to-reader locking” where the tag responds only to the        reader which awakened or “activated” it from a low power        “hibernate” mode, thus improving interference rejection    -   Other features built into the command set and tag state machine        response to provide interference rejection, such as “session        locking”. Previous Class 1 oriented RFID system design assumed a        “promiscuous” tag that responded freely to all reader commands        it heard, with very limited protection from false responses.        According to this disclosure, much more sensitive tags using        this promiscuous mode will commonly mistakenly respond to        commands from enemy readers.    -   Programmable sensitivity upon activation and deactivation    -   Activation selectable as to type of tag (Class 3, Class 3 Plus,        Class 4, tags with sensors, etc), and activation extendable to        control active tag features    -   Reuse of session flags as timer flags in power leveling, and        reader programmable precision timers on the tags for optimum        power leveling control    -   Additional command set features to support power leveling, such        as reader reporting to tag any offset between forward and        reverse reader carrier power (for Class 3 power leveling), or        directly reporting reader forward power (for Class 3 Plus and        Class 4 power leveling)    -   Combining “Select” for choosing tags to enter a query round (tag        singulation and access operation) and “Query” (to actually enter        a query round) command functions into a single command for        faster operation. This may be particularly important for quickly        pulling high value tags such as sensor tags into a query round,        such that these high value tags get priority.    -   As battery tags can be quite sophisticated and have a large        range of features, providing reader awareness of tag        capabilities though on-tag “Capabilities Files” and control of        these features through on-tag “Setting Files”. For example,        programmable duty cycle may control tag normal and hibernate        mode receiver duty cycles.    -   Providing a directory system on the tag (Tag Capabilities and        Reporting System Map, or TCRS) for accessing capabilities and        settings files efficiently, with the minimum number of reads in        the hostile radio environment, and for growth over time of the        TCRS system

FIG. 19 illustrates an exemplary Activation command structure forSemi-Passive Class 3 tags according to various embodiments of theinvention. The fields shown within the command structure may vary inlength and type as dictated by its Extension Flag 1914. The structuremay comprise a preamble 1900, for example one based on PN sequences asdiscussed earlier such as FIG. 13. The Activation Control field 1902allows the reader to command post-activation forward data rate and isextended to also include programming of post activation tag sensitivityaccording to certain embodiments of the invention. This ActivationControl field is further detailed in FIG. 22.

A Target field 1904 (see FIG. 24) identifies tag flag states upon whichactivation may be made selective. But, in contrast with prior art, thetarget field may also include a sub-field specifying category orcategories of tags to be activated, such as Class 3, Class 3 Plus, Class4, tags with sensors, combinations thereof, and additional futureoptions. Prior art has previously included a “Mask” field 1910 that isto be compared against memory or register contents on the tag, which isreferred to as the “Activation Code” AC within ISO/IEC 18000-6C. Whenother requirement of the activation are met, such as required tag class,then a match between Activation Code on the tag and Mask as sent in theActivation command authorizes the tag to awaken. The prior art MaskLength field 1906, Address/Offset field 1908, and Mask field 1910 areused to precisely bit align the transmitted Mask with the desiredportion of the prior art Activation Code stored on the tag. Only tagsthat have a perfect Mask to partial or full AC match will activate.

A “Reader/Interrogator Information” field 1912 (see FIG. 23 for fullexpansion) may also be integrated within an Activation commandstructure. This Reader Info field 1912 may be used to enable thetag-to-reader locking feature that prevents BATs activated by one readerfrom acting upon other commands of other readers. It provides theInterrogator ID needed by the tag to associate with future readercommands, a bit to control whether Tag-to-Reader Locking is to be ineffect, and a Regulatory Region field that informs the tag of thegeographic region of operation. Extension Flag field 1914 informs thetag if there is more information to follow, such as a Class 3 PlusActive Transmit Set Up field, or if the Activation command is about toterminate. The command ends with CRC 1916 which may optionally be usedby tag to confirm almost certain perfection in the received Activationcommand.

The activation and other commands may also be expanded for the moreadvanced tags, taking advantage of the Extension flag. Examples includeextensions to command active transmit and receive channels.

FIG. 20 illustrates an exemplary Activation command structure forSemi-Active Class 3 Plus tags according to various embodiments of theinvention. This command structure is similar to the Class 3 Activationcommand of FIG. 19 and may also wake up Class 3 tags in parallel withClass 3 Plus, depending upon setting in the Target field of FIG. 24. Inthe case that the reader is sending this expanded Activation command,the first Extension Flag 2014 is set and is followed by Active Tx Set upfield 2016 (further described in FIG. 25). This active transmitter setup field is followed by second Extension Flag 2018 which is not setsince the command does not include Class 4 Active Rx Set Up Information.The command ends with the CRC 2020.

FIG. 21 illustrates an exemplary Activation command structure forFully-Active Class 4 tags. This command structure is similar to theClass 3 Plus Activation command of FIG. 20, except with more informationappended. It may also wake up Class 3 and Class 3 Plus tags in parallelwith Class 4 tags, depending upon setting in the Target field of FIG.24. In the case that the reader is sending this further expandedActivation command, the first Extension Flag 2114 is set and is followedby Active Tx Set up field 2116 (further described in FIG. 25). Thisactive transmitter set up field is followed by second Extension Flag2118 which is now set since the command does include the following Class4 Active Rx Set Up field 2120 (further described in FIG. 26). Thecommand ends with the “cyclic redundancy check” or CRC 2122, whichprovides a high reliability error check at the tag that the command wascorrectly received.

FIG. 22 expands the Reader Activation Control field of the variousActivation commands. Reserved for Future Use (RFU) field 2202 allows forfuture change. Activation Version field 2204 allows for selecting theLong Activation command of FIG. 19 or a Short Activation command (notshown). Data Rate field 2206 allows for informing the tag of what readertransmitted forward data rate will be used following activation.Sensitivity field 2208 allows for programming the tag sensitivity to beused in normal mode. This field would most commonly be used to programhigh sensitivity, but in higher interference environments may be used toprogram low sensitivity and thus prevent the tag from responding to orbeing as badly interfered with by more distant readers. However, oneskilled in the art will recognize that this field may be used fornumerous different command features. The field ends with additional RFUbits 2210.

FIG. 23 illustrates an exemplary expanded reader information field forActivation commands according to various embodiments of the invention.This reader information field contains information to be used to enablethe tag-to-reader locking feature to prevent BATs activated by onereader from acting upon other readers' commands. In certain specificexamples, this field includes Reader ID field 2302 with an 8 bit “ShortReader ID,” allows a combination of up to 256 reader ID codes.

Reader Lock field 2304 informs the tag of whether or not to applytag-to-reader locking. For example, this functionality may not bedesired in certain situations such as if the tag is to be deliberatelyaccessed by multiple readers in normal mode. When tag-to-reader lockingis enabled, readers append their IDs in a field to normal mode commandsso that BATs may only respond to commands from the particular readerwhich activated the corresponding BAT. This behavior is important in thecase of the high sensitivity tags disclosed herein in order to limitundesired tag responses. Accordingly, a reader may adjust itstransmitted power to a high power level to reach a desired distantlocked tag while still maintaining a low probability of disrupting othertags that have been engaged by other readers.

In various embodiments of the invention, this expanded readerinformation field may be integrated within the Activation commandstructures shown in FIGS. 19, 20, 21, and 30. FIG. 23 also contains aRegion field 2306 which may inform the tag of geographic region orregulatory region of operation. Different areas of the world havedifferent radio regulatory requirements, and the tag may use the Regionfield information advantageously to adapt its operation. For example, itmay adjust front end RF filtering to optimize to the band in use in acertain region, or observe limits on its transmit frequency and maximumtransmit power that are in keeping with local regulations.

FIG. 24 illustrates an expanded Target field of the Activation commandsthat enables the applicant invented feature of selective BAT activationaccording to tag class 2402. The reader also typically awakens tagsaccording to the state of a particular Session flag 2404. In certainembodiments, the normal Session flag states are re-used as time-outindication states while hibernating, for use in controlling powerleveling. As will be discussed in more detail later, closer range tagsthat have been accessed at a lower reader transmit power level are putback to hibernate with a timer running, and its running state isindicated in hibernate mode by the Inventory Flag being in symboliclogic state “B”. This state is indicated in Inventory flag Target field2408, and may be mapped to either electrical logic state. While thistimer flag is in state B, normal power leveling operation is for thereader to wake up new tags that have not been accessed by waking then upselectively using state A of the Inventory flag. Thus, according to theinvention the utility of the Inventory flag has been expanded from thenormal mode to a key functionality needed in the hibernate mode.

When in normal mode, the Inventory (or Session) flags indicate if a taghas recently been accessed. When in hibernate mode, the Inventory flagindicates if the tag was recently activated and accessed, with thedefinition of “recently” being defined at least partially relative tothe timer programming according to various embodiments of the invention.If the reader wishes to wake up tags regardless of timer state, it mayset Inventory Flag Use field 2406 to “Don't Care”. This may be done, forexample, when the reader has taken too long to complete a power leveling“mini-round” and some earlier tags that were hibernating with timersrunning have timed-out and reset their inventory flags to state A, whileothers are in state B.

The reader may wake up a group of tags or all of the tags within asystem using a Don't Care in Inventory flag state in order to reprogramor “refresh” their hibernate timers. Stateful Hibernate Timeout field2410 provides a 4 bit programmable time value up to 4096 seconds. Thetimer accuracy may be specified to various values or ranges, such asbeing at least +/−20% over the nominal temperature range of −25° C. to+40° C., but it is typically much more accurate. In certain instances,the precision of this timer may be critical to reliable power leveloperations, as it provides the reader with a guaranteed time intervalduring which recently accessed tags will not respond. This allows thereader to reserve its higher transmit powers for a small number of moredistant tags that may be quickly accessed, greatly reducing the timethat the reader uses high power, and thus greatly reducing interference.

Embodiments of the invention also provide for the use of “SessionLocking” (also called “Interference Resistance Mode”) whereby the taglimits its responses to commands featuring session flags that match thesession flag it was activated to. In particular this includes theEPCglobal™ Select command and its overpowering effect on passive tagstate machine operation. In certain embodiments, this feature isspecified during activation by the Inventory Flag Use field 2406 of FIG.24. If this field is set to “Do Care” then the tag not only selectivelyactivates upon the state of the timer controlled Session flag state inhibernate mode, but also alters its post-activation Class 1 statemachine behavior to reject commands that would undesirably alter itsstate by observing “Session Locking”. This double use of the InventoryFlag Use field was chosen by the ISO 18000-6C committee over theproposed method of a more flexible separate one bit “InterferenceResistance Flag” or IRF to command the proposed preferred embodiment of“Session Locking” because it could accomplish most of the primarymission of improved interference resistance and save a bit in theActivation command.

When in the “Session Locking” mode the tag shall obey Select commandflag programming and state machine changes only if the Session of theSelect command as specified in its “Target” field matches that of theActivation Session. Unlike promiscuous Class 1 tag behavior, the Class 3tag with Session Locking ON shall only return to the Ready state inresponse to Select commands where the Select command session matchesthat of the Activation Session. Non-matching Select commands areignored. Furthermore, for the preferred embodiment of tag-to-readerlocking in effect, the still stronger interference resistance isincorporated of the tag only responding to commands if the Short ReaderID field of the command matches that provided by the last Activationcommand.

FIG. 25 shows an exemplary Active Tx Set Up field used for Class 3 Plusand Class 4 tags to control the frequency, power, data rate, andoptionally modulation mode of their active transmitters according tovarious embodiments of the invention. Class 3 Plus tags are preferablyalso able to operate in backscatter transmit mode, thus preserving theiractive transmit operation for situations where it is required, thusimproving battery life. The Base Mode field 2500 indicates if the tag isto use its backscatter transmitter, or its active transmitter, and ifactive, over a low or optionally high power range. New Channel field2502 is a bit field indicating desired carrier frequency in incrementalsteps. For example, the New Channel Field 2502 may provide incrementalsteps of 25 kHz and cover a range of 204.8 MHz, which allows coverage ofall known planned RFID bands. Auto Power Level field 2504 instructs thetag as to whether it should use only the transmit power level commandedby the reader, or calculate its own transmit power level based on itsown measurement of link loss. Transmit Power fields 2506 and 2508 areobeyed by the tag if Auto Power Level field is set to “No.” Data Ratefield 2510 is used for the reader to command the tag reverse active datarate. Modulation Mode field 2512 is used to command the tag's returnmodulation mode, if the tag supports more than one modulation mode.

FIG. 26 shows an exemplary Active Rx Set Up field used for programmingClass 4 tags according to various embodiments of the invention. Suchtags would normally also support Class 3 and Class 3 Plus modes ofoperation, with full active transceiver operation being a part timeoption used only as needed. When the active receive mode is used, it mayadvantageously be duty cycled in order to minimize average powerconsumption. Receiver OFF Time field 2600 is used to program the timeperiod the active receiver is off, and Receiver ON time field 2602 toprogram the time the active receiver is on. When the active receiver isoff, the tag has the option of using a low power Class 3 square lawreceiver. New channel field 2604 sets the receiver listening frequency.

Unlike a square law RFID receiver with broad selectivity limited by itswide band front end RF filter, the optional active receiver can have avery narrow bandwidth generated at baseband (direct conversion receiver)or at an intermediate frequency (superheterodyne receiver) that ishighly efficient at rejecting interference from all transmitters notdirectly on the desired channel. Data Rate field 2606 commands postactivation receive data rate. Modulation Mode field 2608 commands postactivation receive modulation mode, if the receiver supports multiplemodes.

FIG. 27 illustrates an exemplary reader command called “Query Rep” withthe option of transmitting the “short reader ID” field in casetag-to-reader locking is in effect for the particular round according tovarious embodiments of the invention. In certain embodiments of theinvention, the reader locking feature may be set at the time ofactivation for eliminating the possibility of BATs acting upon otherreader's commands. Tags that are not otherwise locked to the reader by aparticular agreed authentication feature, such as an exchanged randomnumber key, may make advantageous use of the Short Reader ID code 2706as a way of authenticating the reader command and avoiding undesirableresponse to other readers. To avoid tags accidentally being activatedand locking to an undesired reader, readers may advantageously employtime coordination, directional antennas, and power leveling. Judicioususe of power leveling, whereby readers use low power to access nearbytags, is an effective way to “zone” tags so that they strongly tend tobe accessed by the desired (normally closest) reader.

FIG. 28 illustrates the Manchester mode Next command according tovarious embodiments of the invention. This command comprises a HibernateSensitivity Control field 2804 to set the tag's hibernate sensitivity toa modest sensitivity state or a high sensitivity state. These twosensitivity states may be advantageously combined with two or moredynamic range states in a high sensitivity square law tag receiver asdescribed herein. This command puts single tags back to sleep using the16 bit random number RN16 generated by the tag and communicatedpreviously to the reader as an authentication step. Because the tagauthenticates the reader command this way, this command does not need toemploy tag-to-reader locking via the Short Reader ID code.

FIG. 29 illustrates the Manchester Deactivate_BAT command according tovarious embodiments of the invention. The purpose of the Deactivate_BATcommand is to send groups of tags back to hibernation. Since it is groupcommand, according to embodiments of the invention, it includes a ShortReader ID code in field 2916 in order to allow tag-to-reader locking. Italso may include Hibernate Sensitivity field 2914 similarly to theimproved Next command. Override field 2912 is another improvement, andallows sending all tags back to hibernate regardless of session. Thisallows a reader, when deliberately intended, to take rapid control andreturn all tags to a known state.

FIG. 30 shows the Class 3 Activation Command extended to provide a PowerInformation field 3014 for enhanced RF power control operationsaccording to various embodiments of the invention. An exemplary PowerInformation field is shown in FIG. 31. In this particular expansion asingle bit 3102 is included as a switch to indicate to the tag if thefollowing 5 bit field 3104 represents current reader forward power orthe change the reader will introduce in its carrier power betweenforward and reverse modes. The switching of field function allows a moreflexible command.

The reason for introducing an offset in reader transmitted forward linkand reverse link power as a preferred embodiment of this invention isthe asymmetric Class 3 link physics as discussed earlier and shown inFIGS. 2 and 3. As illustrated in FIG. 3, over typical Class 3 ranges ofa few meters to a few tens of meters, forward link power may typicallybe from 10 to 40 dB less than reverse link power while still maintaininga matched link condition. This physical fact leads to the advantageoususe of more carrier power in reverse mode than forward mode, whichcauses much less reader on reader interference since the reverse carrieris very spectrally pure and does not cause interference to readerslistening on other channels. But, if this advantageous behavior is used,the tag may need to know the offset between forward and reverse power inorder to adapt its backscatter power for both interference control andregulatory compliance reasons. Because the tag can measure forwardpower, if it is told the reverse power offset from forward power, it maythen easily calculate reverse carrier power as seen at the tag, and usethis information to control its backscatter. Alternatively, the 5 bitfield 3104 may directly indicate the forward power to be used, asspecified by function switching bit 3102. This allows the tag tocalculate the path loss from its measurement of forward power asreceived at the tag. If the reader then uses either no offset fromforward power, or an offset known to the tag from other methods, thenthe tag intelligently adjusts its backscatter from this informationalso.

FIG. 32 is an exemplary expansion of a reader power information fieldthat may be used in the case of Class 3 Plus and Class 4 tags for apower leveling enhanced Activation Command similar to FIG. 30, butextended to Class 3 Plus and Class 4. In the case of the Class 3 Plusand Class 4 system, there is no reader reverse mode carrier, and insteadcontrol is applied to the active carrier generated by the tag. In thiscase there are two 5 bit fields, one to indicate forward reader powerand one to indicate desired reverse power, or reverse power change froman assumed nominal tag transmit power, or reverse power as desired to beseen in the reader receiver. Functional switching fields may be added tothis field to allow all these options. This field could also besimplified down to be only the reverse mode active carrier information,if the Power Information field of FIG. 30 were also included in anActivation Command with improved power leveling capability that coveredClass 3, Class 3 Plus, and Class 4.

FIG. 33 shows the new Flex Query command according to variousembodiments of the invention. This command corrects a situation in whichpreviously required separate Select and Query commands are required inorder to selectively bring different categories of tags (such as ID onlytags and sensor enabled tags) into interrogation/query rounds. Inpassive systems in particular this is problematic, as the lowsensitivity of passive tags lead to tags in motion only having briefwindows of time in which they can be accessed. This new command allowsfaster access while still maintaining the ability to selectively bringtags into Query rounds based on their basic types and attributes. Thisis referred to herein as a “mini-select” function, as compared to a fullfeatured separate “Select.”

The types of tags are selected in the Tag Type Select field, which isdetailed in FIG. 34 according to various embodiments of the invention.The 12 bits of this field allow selection of any combination of PassiveClass 1 and 2, Semi-Passive Class 3, Semi-Active Class 3 Plus, severalvariation of Class 4 (such as simpler Class 4 that do not featuretag-to-tag networking, and more advanced Class 4 that may featuretag-to-tag networking), several types of sensors including so calledSimple and Full Function sensors, tags with sensors that haveexperienced an alarm condition (such as temperature out of range), andRFU bits for future expansion.

Certain embodiments of the invention provide the Flex Query an abilityto control Simple Sensor entry into interrogation/query rounds and alsocontrol of Simple Sensor response. “Simple Sensors” are defined assensors with a set of preprogrammed behaviors that generate a smallamount of sensor data (for example, a notification that a temperaturesensing tag has been exposed to temperature limit outside itspreprogrammed range) and that were originally intended to automaticallytransmit that data in addition to their identifying data when the tagwas properly singulated. Simple Sensors automatic response allows theirdata to be read by the reader without taking the time to choose SimpleSensor tags via the “Select” command for inclusion in a query round.Taking such time, particularly in the case of tags with less thanexcellent sensitivity, leads to a statistical increase in the tag readfailure rate. But, having Simple Sensors always transmitting their data,in case of many Simple Sensor tags mixed with a population of tagswithout Simple Sensors, would noticeably slow the singulation process,also statistically leading to an increase in the number of missed tagsthat were not properly singulated and read.

Embodiments of the invention build the “mini-select” functionalitydescribed above into a specialized query command that could then choosewhether or not Simple Sensor commands would enter the query round, andalso whether their automatic response function would be in effect forthe particular query round.

In further embodiments of the invention, a system is provided of TagCapabilities Reporting and Setting (TCRS) for reporting and controllingcapabilities of advanced tags. Battery tags can be very sophisticatedwireless terminals with many options, and the TCRS system provides aconvenient and adaptable means for the tag to report its possiblycomplex capabilities, and for the interrogator to command behavioralmodes. An option under TCRS is to allow interrogator control of dutycycle to optimize battery power and tag latency of response.

Among capabilities to report, there are numerous possible combinationsof Simple PIE, Advanced PIE, Manchester, Dead Battery Response (wherethe Class 3 tag with a dead battery can act as a passive tag), andPassive Fall Back (where the tag with Class 3 receiver completely offcan use a passive receiver) that may be relevant. There is also a widerange of tag sensitivities that can exist and which may be provided tothe reader for system control, from approximately −80 dBm (sensitivesquare law receiver plus RF low noise amplifier) to approximately −15dBm for passive tags or passive mode receiving on a BAT.

The TCRS system may use Capabilities Files and Setting Files in tagmemory, where the detailed file definitions are linked to a VersionCode.

FIG. 35 illustrates the memory structure of an initial Tag Capabilitiesand Reporting system according to various embodiments of the invention.This constitutes a simple but effective file system for managing thisinformation. An initial pointer to the TCRS Map (directory), here TCRSMap Address 3504 and 3506, is placed in a known memory location where itis not subject to being written over, such a standardized location inTag ID or TID memory. In addition to this address pointer, a TCRS MapSize 3502 is included so that the reader can access the full TCRS mapand no more in a single efficient read operation. This pointer points tothe first word of the TCRS Map, here Version Code word 3530. In certainexamples, the TCRS Map may be stored in the User memory bank of an ISO/IEC 18000-6C tag.

The TCRS Map also comprises a status word 3528. Above that are threewords specifying the size of the User memory, followed by Block Sizeindicator word 3520. The block size is used in certain memory accesscommands. For example, TCRS Map Entry 1 for Capabilities 3518 points tothe address of the Capabilities File 3512, and provides information asto the used and maximum allowed size of this file. TCRS Map Entry 2 3516provides similar directory information for the Settings File 3514. Thesingle read of the TCRS Map may also provide the version and directoryinformation needed for the reader to access Tag Capabilities File 3512and Tag Settings File 3514. In certain embodiments, the Tag Capabilitiesfile is stored in “locked” memory for security, and the Tag SettingsFile will be in “unlocked” memory for reader updating.

FIG. 36A provides an example of a Battery Capabilities Word (BCW) in aCapabilities File according to various embodiments of the invention.This illustration provides a small fraction of the myriad possibleoperating capabilities that the tag may have and that the reader may beaware of.

FIG. 36B provides an example of a Battery Settings Word (BSW) that maybe part of a Settings File according to various embodiments of theinvention. This particular word shows the feature of tag receiver dutycycling. Since RFID tags spend most of their time listening and nottransmitting, the listen mode power consumption is critical. In general,better sensitivity in tag receivers requires more power consumption, sothere is an inherent trade-off between sensitivity and battery life. Oneway to optimize this operation is to “duty cycle” the tag wherein itperiodically listens for some period of time, the shuts down to savepower. This repeats at a rate suitable to maintain low latency. To bestserve various applications the duty cycling may be adjustable. Apreferred way to control this adjustment is via the Battery SettingsWords disclosed herein.

N. RF Power and Interference Control in the Class 3, Class 3 Plus, andClass 4 Cases

“Power leveling” is a term for transmit power control that has itsorigins in the cellular industry, where a group of handset transmitsignals as viewed on a spectrum analyzer display at the base stationhave similar received power or “level”. Powers that are “level” towithin the transmitted adjacent channel splatter of the handsets keepsthe splatter from one handset from unacceptably interfering one channelover, allowing the base station to keep handset signals separated byfrequency filtering. By also keeping power as low as is consistent withreliable communications, interference is geographically limited,allowing frequency reuse at shorter ranges and thus maximizing use ofthe radio spectrum. Lower powers also greatly reduce the creation ofhigher order, particularly 3^(rd) order, intermodulation products.Because 3^(rd) order products are “in-channel” they cannot be filteredoff. Since they grow proportional to the 3^(rd) power of the drivingsources (3 dB per dB), they drop rapidly as power is reduced and thusare much more manageable for lower receive powers. Hence, power levelinghas become standard practice in dense wireless systems, which are bytheir nature “interference limited”.

In RFID, the reader is comparable to the base station of a cellularsystem, but it does not intentionally simultaneously listen to multipletags at the same time whose power must be “leveled” as seen at thereader. Hence the term “power leveling” is not exactly analogous. Whatis really needed in RFID is “power control”, and particularly of thereaders in order to prevent reader on reader and reader on taginterference. However, in this disclosure the phrase “power leveling” isused as a pseudonym for “power control” since it is such a commonwireless industry term.

Various embodiments of the invention provide variations of power controlsuited to the asymmetric link physics of backscatter RFID. In the caseof battery tags, this physics leads to the need for ultra-sensitivereaders, and this in turn requires strict power control on the part ofother readers in order to control the interference they impose both bysplatter and by intermodulation product creation in victim readers. Thewideband nature of the direct detector receiver in the tag, whichnormally simultaneously listens to the entire regulatory band allocatedto the RFID system, is also prone to high interference as the tagbecomes more sensitive. Thus, there is a compelling need for powercontrol in these RFID systems.

Reader-on-reader interference is particularly troublesome inSemi-Passive Class 3 in forward mode because the readers are extrasensitive as compared to passive systems. Co-channel and even adjacentchannel transmission at higher powers can cause interference at greatdistances. For this reason “split band plans” that separate the forwardand reverse links into separate band segments with sufficient safetyband of unused frequency between them for good filtering at the readerwould be of great benefit. A particular troublesome case of readersinterfering with other readers occurs when they are trying to listen toweak tag backscatter. Advantageously, in the split band plan case theinterference from readers nearby in frequency is not spread out byforward modulation (only quiet carriers to support backscatter are nearin frequency) where it overlays desired tag sidebands. Thus, in the caseof split band plans the selectivity of the reader may be used to rejectthe nearby in frequency enemy reader carriers.

In the absence of such split band plans, the use of power leveling iseven more important, since an enemy reader may be in the very nextchannel to the one where a reader is listening to low level tag replies,and if it is at high power its splatter may easily interfere with tagbackscatter. Based on the extended command set disclosed herein,advantageous methods of power level control may be implemented.

According to various embodiments of the invention, the use of precisionprogrammable timers in the tags are used relative to RFID power controland the use of intelligently controlled and sometimes differing forwardand reverse reader powers are employed. Both of these are extensions ofthe power control within the practice of RFID and certain embodimentsmay be explained referring to FIGS. 2, 3, 37 and 38.

Reviewing FIG. 2, it is apparent to one of skill in the art that thereis a large difference in required tag and reader sensitivity due to theasymmetric link physics of backscatter systems. When tags are extendedto sensitivities of −40 dBm and below, the limits of noise physics donot allow the reader to maintain a matched link in the case of equalforward and reverse reader power. At a tag receive sensitivity of −40dBm, attained with the square law receiver methods according toembodiments of the invention, the reader would have to attain and use asensitivity of −110 dBm to keep up. This is right at the limit ofphysics and is very difficult to attain. As an example, a trulyoutstanding reader in current technology might attain −105 dBm relativeto total received backscatter counting backscatter carrier andsidebands. Lower levels corresponding to tag sensitivities much below−40 dBm are not attainable at practical data rates.

To maintain system performance it may become necessary for the reader toactually be able to use the full sensitivity that it does attain. Thusfor Class 3 systems interference as seen at the reader at the frequencywhere tag sidebands lie preferably should have a level constrained to beabout −110 to −120 dBm and below. Interference as seen at the taganywhere in the band during the time the tag is receiving should be onthe order of at least 10 dB below the desired reader signal as seen atthe tag, and preferably at a general level below about −50 to −70 dBm.Higher levels can be temporarily allowed through time coordination andthrough locally higher desired powers that overcome interference levels,but in accordance with the material presented in this application, theneed for low interference levels is clear.

A review of FIG. 3 reveals the opportunity for reduced forwardtransmission power levels as compared to reverse backscatter supportingcarrier power levels. At typical Class 3 RFID ranges of a few meters toa few tens of meters, this difference can be approximately 10 to 40 dB.Reducing forward carrier power several tens of dB on average allows formuch lower interference levels as seen at the reader due to spectralsplatter than can overlay tag sidebands as seen in the reader. Statedanother way, the backscatter supporting reverse link power shouldgenerally be higher than the forward link power by about 10 to 40 dB.Due to the non-linear nature of intermodulation product creation, itreduces intermodulation product interference even more. For example, a20 dB reduction in 3^(rd) order intermodulation creating powers reducesthe created 3^(rd) order products by 60 dB.

FIG. 37 illustrates the concept of “mini-rounds” used to discover andcommunicate with a plurality of tags according to various embodiments ofthe invention. The mini-round concept is actually applicable to Class 1,Class 2, Class 3, Class 3 Plus, Class 4, and mixed class system. Herethree mini-rounds 3702, 3704, and 3706 are depicted, but circumstancesof a particular system may lead to a different number. Geometricallytags in a logical mini-round tend to be in band between a minimum rangeand a maximum range suited to the forward power level in use in aparticular mini-round.

The general method of working with mini-rounds is for closer tags to beaccessed at lower reader transmit powers, and then put to sleep withprecision reader controlled timers running that prevent their respondingto Activation commands aimed at more distant tags using higher readerpower. When accessing more distant mini-round tags, the reader hasalready accessed closer tags at a lower power, and is not usingsufficient power to reach beyond a certain maximum range. The timervalues chosen by the reader are based upon the expected number of tags,the data rates in use, the amount of information to be read from thetags, the interference suffered, any necessity for the reader to refrainfrom access due to time sharing of the radio band with other readers,and other factors. However, the time is picked with the intent that thereader can conduct a full set of mini-rounds out to a maximum desiredrange before the inner ring tags time out and begin responding.

In certain embodiments of the invention, each mini-round is a separatelogical Query round, and within each such mini-round/Query round theremay be separate “quasi-rounds” used with the “QueryAdjust” command inorder to resolve tag collisions from tags that use the same time slot.The general method of tag singulation, as the tags have only widebandreceivers, is to separate tag replies in time using random time slots.This wastes some time for slots where no tags reply, but allows the keycost and power reducing factor of wideband tag receivers. By using asufficiently large number of random time slots relative to the tagpopulation, most tags are read without collision. The small number oftags that do collide are then efficiently read in a follow up“quasi-round” using a smaller number of random time slots suited to thenumber of tags that actually collided.

If the reader miscalculates the time needed and inner mini-ring tags areabout to time out, the reader may revert to low power, awaken those tagsby pointing to state B in the Activation command (the state indicatingthat the timer is running), and reprogram or “refresh” the timers. Thetags may then be put back to sleep at the same time with Deactivate_BATwithout being accessed. If some but not all tag timers in a mini-roundhave expired, the reader may use a “Don't Care” on Timer/Session flagstate. All tags in range will wake up, possibly including some that arebetter associated with other readers, and all these will have theirtimers associated with the activation session reprogrammed. However,since there are independent timers available for each of the foursession flags, this accidental programming is not a serious problem.

In certain embodiments of the invention, readers may broadcastinformation about the forward and reverse powers to enable BATs toadjust their backscatter power, effectively enabling reverse link powercontrol. There may be certain systems that are required to limit tagemissions, such as those deployed in Europe which has enactedregulations limiting the amount of tag emissions. Information aboutreader transmit power in combination with measured link loss in the tagallows the tag to meet these limitations by appropriately controllingthe fraction of maximum available backscatter that it reflects.

In a “matched” receiving state, a tag antenna has a load impedance equalto the antenna impedance (which is mostly radiation resistance for anefficient antenna). While receiving maximum available power, the antennawill also reradiate some amount of power. In the “matched” case whereantenna impedance is correctly conjugately matched to load impedance,equal current actually flows in both the antenna and the load, and theantenna actually reradiates (backscatters) just as much power as isdelivered to the load. Maximum possible backscatter is theoreticallyfour times larger than the maximum receive power at the tag for a givenfield due to the fact that if a short is switched across the tagantenna, then the current flow induced in the antenna doubles sincetotal impedance limiting current flow is cut in half. As such, there isno power delivered to the zero impedance “load,” (since power=i²R, and“R” is zero) and power backscattered goes up as the square of thecurrent. If this power is found to be in violation of regulations forthe measured receive power, then the tag may calculate how much itshould degrade or limit the backscatter in order to meet the regulatoryrequirements, because the tag is aware of how much reader power will goup when the reader transitions from its own transmit to receive state.As described earlier, this information may be coded in a field incommands that precede tag transmissions. The Class 3 tags can use theinformation on the difference between forward and reverse power in orderto be aware of the carrier power that the reader sends in reverse mode,in order to in turn control their emissions based on reflectivebackscatter. This control may advantageously be in a precise closed loopor feedback form. Such a feedback loop may be analog or digital, or acombination.

The Class 3 tag does not have to be aware of the actual link loss tocontrol its emissions, as it generally simply needs to know what thereverse carrier is at its operating range so that it does not overreflect that available power and thus exceed desired tag emissions. Itcan know the available power for backscattering by measuring thereceived forward power and understanding the difference between forwardand reverse reader power in a variety of ways. These powers could be thesame, have a fixed difference, a temporarily fixed difference, or adifference that the reader dynamically informs the tag of in activationand other commands. The reader can also fine tune the backscattersupporting carrier power it sends. For example, if the reader sensesthat backscatter power from a particular tag is more than it needs, itmay in its communication with that particular tag reduce its backscatterpower with or without informing the tag. This has the additional benefitof reducing reader on tag interference to other tags that may be tryingto hear their own readers in forward mode at this same time. Recall thatsince the tag receiver is wideband, it has little rejection of “enemy”reader transmissions.

The reader power preferably also varies with the interference situation,and with the performance of the particular reader. For example, if thehighest 10 dB of reader transmit carrier power desensitizes the readerreceiver due to carrier leakage and phase noise effects, then the onlyreason for the reader to use reverse carriers in the highest 10 dB is toovercome interference. That reader should then preferably only use thehighest 10 dB of its own backscatter supporting carrier if it senses itneeds to overcome interference. This amount of self desensitizing alsovaries with subcarrier and data rate in use, so may be intelligentlyadapted as a function of these parameters. A well designed Class 3reader may achieve full sensitivity at full power with largersubcarriers, but generally not with smaller subcarriers.

The reader may adjust the data rates used in each mini-round in additionto adjusting its power level. In the reverse link limited instance oflonger range Semi-Passive Class 3 operations, a lower tag data rateallows a reader to adjust its channel bandwidth to a lower value, thusreducing its noise floor and allowing improved SNR, and thus successfulcommunication.

Class 3 Plus and Class 4 tags need the actual link loss so that they maytailor their transmit power to overcome that link loss. Class 3 Plus andClass 4 tags can calculate such link loss if they are informed by thereader of its forward link power, as they can measure the receive powerthey see from the reader, with the difference between transmit readerpower and that received at the tag being the link loss. They may thenset their transmit power to deliver a desired receive power at thereader. The reader may instruct them to adapt that power as necessary tohave sufficient receive power, but not an excess of wasted power thanonly increases interference. Class 3 Plus and Class 4 power levelingwill be covered in detail towards the end of this section.

FIG. 38 is a flowchart illustrating an exemplary forward power controlmethods according to various embodiments of the invention, particularlyfor Class 3 systems. The Class 3 power leveling also applies to Class 3Plus and Class 4 tags operating in their Class 3 “fallback” mode, whichthey may be expected to do a significant fraction of the time in orderto maximize their battery life.

This particular example uses the flag names defined in EPCglobal™ Gen 2standard version 1.2.0 with the Class 3 extensions of ISO/IEC 18000-6C,and assumes a mini-round based system. One skilled in the art willrecognize that other flag naming protocols may be used as well asvariations to the mini-round discovery operations.

When power leveling in Class 3 systems, a reader may send an Activationcommand at a low power P₁ over a set of powers P_(i) 3804, optionallyusing the tag-to-reader locking and/or session locking features, whichwakes up nearby tags of the desired type and inventory flag state. Asthe tags are awakened their timers are programmed for a time that thereader judges adequate to complete a full set of mini-rounds. The readerthen issues Select command 3810 (if desired), though with the disclosedmethod of a “mini-select” built into the Query_BAT commands for bothAdvanced PIE and Manchester, the Select command is often not needed. Thereader next sends the Query_BAT command 3812 for the reader's assignedsession, still using lowest forward transmit power P₁. All tags matchingthe Query criteria enter into the mini-round.

Modern UHF RFID systems based on EPCglobal™ Gen 2 standard version 1.2.0separate tags in time in a logical Query round (each mini-round) by useof random numbers generated in the tag and loaded into a counter that isdecremented by a reader command (the QueryRep command). As tags count tozero, they may be generally individually accessed (singulated) becausethe random counter was selected large enough to allow for reliablyseparating most of the tags in time. Those that “collide” on the samerandom “slot” start over with their counter at a high value that wouldgenerally never be counted down to zero, simply expecting to later enteranother “quasi-Query round” via the QueryAdjust command where they get anew and smaller random number to count down. When tags reach countervalue zero, they reply 3816, and read and/or write operations areindividually conducted. At the conclusion of reader access, the readersends a QueryRep 3814 or QueryAdjust 3830 to get to the next tag, andthe Inventory flag of the just inventoried tag changes state. In otherwords, the state changes to logical state “B”, meaning it has beenaccessed and should not reply until reset to “A” by either a “Select”command. Generally it will not be accessed again, simply put to sleepwith timer running and not accessed again until after the timer expires.

On this same QueryRep 3814, another tag may reply with its RN16 or otheridentifier which is noted by the reader. But, before the reader dealswith the new tag, it first puts the tag it last accessed back to thehibernate or sleep state by sending the Next 3824 command with the RN16handle of the previous tag. The last tag then goes to the hibernate modewith its inventory flag set to B, and it stays in B until timing out oruntil awoken and reprogrammed via a new Activation command thatdeliberately activates on timer/inventory flag state A (normally onlydone if “refreshing” of the timer is needed). Note that the Next commandis generally preferred to put tags back to sleep if there have been anycollisions, as use of Deactivate_BAT would also send the tags thatcollided and were not accessed back to sleep. Since the reader is notalways sure if there have been collisions, and cannot know if there willbe collisions as the mini-round progresses, Deactivate_BAT is in generalreserved for low tag density situation and the timer “refresh” operationdescribed above under the discussion of FIG. 37.

When the reader has stepped through the 2^(Q)−1 QueryReps 3814 neededfor this “mini-round,” it repeats with QueryAdjust 3830 and a smaller Qif there were collisions 3818 leaving a few tags left unread in thatparticular mini-round.

When there are no collisions 3818, the reader is ready to step up to thenext higher power level P₂, and issues a new Activate command 3808, thuswaking up more distant tags of the desired type. The tags just accessedare all in timer state B in hibernate and do not respond to this nexthigher power Activate command. The inventory process is then repeatedwith this new set of tags. The reader knows the timeout period andearlier low power level of the tags it has put back to hibernate, and ifthe total length of the round approaches the flag reset time it revertsto low power, wakes those tags back up with inventory flag state set totheir current state or “don't care”, and resets their timers. It thenreverts to higher power and completes the round.

This process is repeated over however many mini-rounds are appropriatefor the local system design and interference environment. In general, alarger number of mini-rounds reduces interference because there are thenonly a small number of tags that are accessed at higher reader powers. Asmall number of tags accessed at higher power equates to a small timebeing used for high power operations, thus statistically reducing theincidence of interference.

Next is covered the special case of Class 3 Plus and Class 4 when theyare using their active transmit capability. The active transmitcapability of these modes relieves the greatest weakness of the Class 3system, which is the reverse link limit and sometimes impossibility ofthe reader attaining adequate sensitivity to maintain a matched linkwith a sensitive square law tag. Even if the reader does attainoutstanding sensitivity, it may still be subject to interference in somecases where its full sensitivity may not be taken advantage of. Thepart-time active transmit capability of the Class 3 Plus and Class 4system can lift the effective transmit power of the tags by many tens ofdB's, and do so within the limits of a small battery such as a lithiumcoin cell. This lifts the receive power as seen at the reader to be wellabove its sensitivity level, and in well controlled systemimplementations well above the interference level as well. The Class 4tag has the additional advantage of part time higher sensitivity in thetag, and far more selectivity, than a square law receiver can provide,while still maintaining square law receive capability for maximumbattery life.

However, active RFID still requires transmit power level control.Furthermore, when fully integrated with semi-passive Class 3 asdisclosed herein, the active tag transmit levels can cause significantinterference to Class 3 operations. The power leveling used in theactive modes needs full integration with the hibernate mode of Class 3for maximum total system performance.

A preferred embodiment to achieve these goals is to design the powerleveling into the activation and normal mode command sets, whereby powerleveling commences in activation and gracefully continues across thewake-up to normal mode operations. This is achieved by providing theClass 3 Plus and Class 4 tags with not only the extensible activationcommand to control their active circuitry, but also with the key readertransmit and receive information needed for power control as part of theactivation command. These tags can measure their own receive power inhibernation or quasi-hibernation (a low power but not fully activatedstate that may occur after signal detection and passing a test such asPN sequence correlation as presented earlier) before full activation. Ifthey are then informed in the activation process of the reader power,they may calculate path loss. They may then calculate their own desiredtransmit power in order to present a desired received power as seen bythe reader. They may also be instructed by the reader to use a giventransmit power that the reader system control software chooses becauseof the need to control tag to reader range, and/or to place tag transmitpowers at desired levels as compared to interference.

This process of informing the tag of necessary link parameterspreferably begins in activation and continues to most normal modecommands. Every command received by the tag is an opportunity to adjustthe link parameters so as to track path loss variation due to multi-pathfade, or to adjust relative to changing interference, without usingexcessive or unnecessarily high powers that cause interference. It isthe best band citizenship possible.

In certain embodiments several variations of tag transmit power controlare provided. For example, to limit tag to reader range the tag may beinstructed to transmit a given power level that the reader instructs itto. This level varies with desired range, data rate, reader sensitivity,interference, and other factors. Alternately, the reader may instructthe tag to calculate its own transmit power based on desired receivepower at the reader, a parameter that may be communicated to the tag inthe fields of FIG. 32. From either an initially commanded tag transmitpower or one the tag calculates based on information provided by thereader, the reader may on every command adjust the tag transmit power,either in absolute terms (such as transmit “X” dBm) or relative terms(such as adjust last transmit power up or down “Y” dB). The inherenthigh level of reader control over the tags with a high density of readercommands under EPC Gen 2 derived protocols is ideal for managing tagtransmit power on a dynamic basis.

While the invention is susceptible to various modifications andalternative forms, a specific example thereof has been shown in thedrawings and is herein described in detail. It should be understood,however, that the invention is not to be limited to the particular formdisclosed, but to the contrary, the invention is to cover allmodifications, equivalents, and alternatives falling within the spiritand scope of the appended claims.

1. An RFID tag configured to receive an RFID Activation commandstructure that is transmitted from an RFID reader and processed in theRFID tag, the RFID tag comprising: an antenna; circuitry coupled to theantenna, the circuitry being configured to process the commandstructure; the command structure comprising: a plurality of fields, theplurality of fields comprising: a session/inventory flag identificationfield that specifies a session that the RFID reader will use after theRFID tag has been activated; a field that indicates whether the RFID tagis to check a hibernate mode state of a session/inventory flag as aparameter upon which to base activation of the RFID tag; a control fieldthat specifies if tag session locking is to be in effect afteractivation of the RFID tag; and a value that places the RFID tag in atag session locked mode to prevent a response to a normal mode commandwith a differing session code than one indicated in the Activationcommand structure, wherein the session/inventory flag state iscontrolled by an RFID reader programmable timer while in a hibernatemode, wherein an activation of the RFID tag based on the timercontrolled session/inventory flag state is related to system transmitpower control operations such that the RFID tag accessed at a lowerreader transmit power level temporarily does not respond to readercommands transmitted at a higher power level.
 2. The RFID tag of claim 1wherein the command structure further comprises a reader lock field thatindicates if the RFID tag is operable to utilizes tag-to-reader lockingwith the first RFID reader after the at least one RFID tag has beenactivated.
 3. The RFID tag of claim 1 wherein the command structurefurther comprises at least one duty cycle field, within the plurality offields, that specifies a duty cycle of operation of a receiver withinthe activated RFID tag.
 4. The RFID tag of claim 1 wherein a statemachine operation is altered when the RFID tag is locked onto a firstRFID reader, the altered state machine operation causing the activatedRFID tag to ignore a command from a second RFID reader that does notfeature an appropriate identification in the reader identification codefield.
 5. The RFID tag of claim 4 wherein a specialized reader commandis provided that overrides the lock between the RFID tag and the firstRFID reader and returns the RFID tag to a hibernate mode.
 6. The RFIDtag of claim 1 wherein the RFID tag is configured to check a hibernatemode tag-to-reader locking flag to identify whether the RFID tag locksto the first RFID reader while in the hibernate mode.
 7. The RFID tag ofclaim 6 wherein the hibernate mode tag-to-reader locking flag is thesame flag as a normal mode tag-to-reader locking flag.
 8. The RFID tagof claim 1 wherein a tag state machine operation is altered from apassive tag operation mode into an interference rejecting mode when atleast one RFID tag is in a session locked mode with a first RFID reader,the altered state machine operation causing the at least one RFID tag toignore a command from a second RFID reader that does not feature anappropriate identification in a session field of said command.
 9. TheRFID tag of claim 8 wherein a specialized reader command is providedthat overrides the locked tag session between the at least one RFID tagand the first RFID reader and returns the at least one RFID tag to ahibernate mode.
 10. The RFID tag of claim 1 wherein the commandstructure further comprises at least one duty cycle field, within theplurality of fields, that describes a subsequent duty cycle of operationof the RFID tag when the RFID tag is activated.
 11. An RFID readerconfigured to generate an RFID Activation command structure that istransmitted from the REED reader to an RFID tag, the RFID tagcomprising: an antenna; circuitry coupled to the antenna, the circuitrybeing configured to generate the command structure; the commandstructure comprising: a plurality of fields, the plurality of fieldscomprising: a session/inventory flag identification field that specifiesa session that the RFID reader will use after the set of RFID tags hasbeen activated; a field that indicates if at least one tag, within theset of RFID tags, is to check a hibernate mode state of asession/inventory flag as a parameter upon which to base activation ofthe at least one RFID tag; a control field that specifies if tag sessionlocking is to be in effect after activation of the at least one RFIDtag; and a value that places the at least one RFID tag in a tag sessionlocked mode to prevent a response to a normal mode command with adiffering session code than one indicated in the Activation commandstructure, wherein the session/inventory flag state is controlled by anRFID reader programmable tinier while in a hibernate mode, wherein anactivation of the set of RFID tags based on the timer controlledsession/inventory flag state is related to system transmit power controloperations such that a subset of RFID tags that were accessed at lowerreader transmit power levels temporarily do not respond to readercommands transmitted at higher power levels.
 12. The RFID reader ofclaim 11 wherein a tag state machine operation is altered from a passivetag operation mode into an interference rejecting mode when at least oneRFID tag is in a session locked mode with a first RFID reader, thealtered state machine operation causing the at least one RFID tag toignore a command from a second RFID reader that does not feature anappropriate identification in a session field of said command.
 13. TheRFID reader of claim 11 wherein a specialized reader command is providedthat overrides the locked tag session between the at least one RFID tagand the first RFID reader and returns the at least one RFID tag to ahibernate mode.
 14. The RFID reader of claim 11 wherein the commandstructure further comprises at least one duty cycle field, within theplurality of fields, that describes a subsequent duty cycle of operationof at least one receiver within the set of activated RFID tags.